Silent data conversion system with sampling during electrical silence

ABSTRACT

An electronic method and apparatus for signal encoding and decoding to provide ultra low distortion reproduction of analog signals, while remaining compatible with industry standardized signal playback apparatus not incorporating the decoding features of the invention, and wherein the improved system provides an interplay of gain, slew rate and wave synthesis operations to reduce signal distortions and improve apparent resolution, all under the control of concealed control codes for triggering appropriate decoding signal reconstruction compensation complementing the signal analysis made during encoding. In addition, signals lacking the encoding process features of the invention are likewise compatible with playback decoders which do embody the invention, to provide some overall restoration enhancement.

This is a division of application Ser. No. 08/110,335, filed Aug. 20,1993, now U.S. Pat. No. 5,479,168, which is a continuation of Ser. No.07/937,631, filed Oct. 6, 1992, (abandoned), which is a continuation ofSer. No. 07/702,073 filed May 29, 1991, (abandoned).

BACKGROUND OF THE INVENTION

This invention relates generally to improvements in signalencoding/decoding methods and apparatus and, more particularly, to a newand improved digital encoding and decoding system for lower distortion,higher resolution, and increased dynamic range reproduction of analogsignals while remaining compatible with industry standardized signalplayback apparatus and standards not incorporating the decoding featuresof the present invention. In addition, recordings lacking the encodingprocess features of the invention are likewise compatible with playbackdecoders which do embody the invention, and are provided someenhancement.

Quite often a recording or communications system is standardized and itsformat cannot be readily altered without affecting a substantialquantity of equipment already in existence. Hence, adding informationwith supplemental codes may not always be practical unless provisionshave been standardized for such insertions. Unfortunately, moderndigital systems are not very expandable since data bandwidth,resolution, error correction, synchronization, ancillary data and other"housekeeping" information essentially occupy the entire digitalcapacity of the storage or transmission medium.

However, electronic equipment manufacturers and users of such devicescontinue to seek enhanced performance and more features from suchstandardized systems. An important example is the need to make acompatible recording well suited simultaneously for portable,automotive, television and audiophile markets. Today, many recordingsare made for the most profitable market while other users suffercompromised sonics. The obvious conflicting performance requirements ofdifferent listening environments and the need for sonic improvementshould desirably be implemented by a new system which is compatible witholder systems and recordings.

Automobile and portable equipment are usually low cost and must operatein noisy environments. Hence, in such situations, a slightly restricteddynamic range playback is beneficial. Audiophile systems require utmostaccuracy, dynamic range, and resolution beyond that which is availablein the current standards. Thus, in any new compatible system, asprovided by the present invention, encoded dynamics and slew ratemodifications which achieve lowest distortion and best resolution forthe audiophile when decided, should also provide improved sonics forportable and automotive playback when not decoded.

Compact Disc pulse code modulation and other digital audio encodingschemes are good examples of highly developed and standardized systemswhich push signal conditioning and digital information limits. Most suchdigital systems originally evolved around then practical 2.5 to 3.5 mHzrotary head video recorder bandwidths. In such standards, the data bitswith error correction and housekeeping entirely fill the availablebandwidth. Accordingly, the need for a "smart" optimization technique,which does not rely upon increased bandwidth for its implementation,becomes apparent.

By way of background, let us consider a typical digital audiorecord-play system, its most frequently encountered components,operation, and difficulties. In its simplest form, the recorder includesa sampling switch and an analog to digital converter. The switch breaksthe continuous analog signal into a series of voltage steps, each ofwhich is converted to number groups or digital words. Digital levelmeters and simple communication systems often operate with just thesefunctions in a single IC chip. Practical high performance record andplayback systems require many added operations to prevent undesiredinternal and external analog-digital signal interactions, as well asbeats and non-linear feed-through between digital and analogfrequencies. Well-known technologies to deal with these problems includesharp cut-off or "brick wall" low-pass filters, fast sample and holdcircuits, and high common mode rejection amplifiers. Unfortunately,although these components and subsystems solve many problems, they alsocreate others.

Briefly, in typical digital recording systems, low-pass filters ring,and if of analog construction, have pre-echo, are subject to suddenphase shifts near band edge, and have capacitors which often causetroublesome dielectric hysteresis effects. Sample and hold circuits haveunpredictable timing and capture errors for different signal slew ratesand also suffer from capacitor problems. Fast digital signals and thehigh speed amplifiers needed to handle them often create and aresensitive to ground currents which can cause audible strobe-beateffects. Digital reproducing systems have similar problems, along withspike or glitch generation caused by digital to analog conversion, anddigital filter word length round off problems. Usually the recorder isdesigned to have state of the art performance while that of thereproducer degrades depending on the economies of "consumer"construction. These and other problems continue to plague modern highperformance digital audio systems.

Unfortunately, such technical difficulties usually create jarringnon-harmonic distortions, typically centered in the most sensitive andperceptive human hearing range. Often these distortions are caused bythe highest, almost inaudible frequencies contained within the programmaterial. Taking the ratio of high and low frequency hearing acuity intoaccount, and the fact that sounds unrelated to the program materialstand out, the presence of even an extraordinarily small amount of thesedistortions can be quite objectionable to the listener. Fortunately,often only very small corrections are needed to minimize some of thesedistortions. However, left as is, these distortion errors can combine toyield the equivalent of 13 to 14 bit performance accuracy from systemsoriginally designed for 16 bit resolution. In practice, while some feelthe advantages of current digital recordings outweigh the disadvantagesof their distortion errors, many sophisticated listeners and audiophilesare not so tolerant.

Accordingly, those concerned with the development and use of digitalsignal encoding and decoding systems for analog signals have longrecognized the need for a higher quality, lower distortion digitalsystem for reproduction of such analog signals, which for all practicalpurposes is also compatible with existing equipment standards. Thepresent invention fulfills all of these needs.

SUMMARY OF THE INVENTION

Briefly, and in general terms, the present invention provides new andimproved digital encoding/decoding methods and apparatus for ultra lowdistortion reproduction of analog signals which are also compatible withindustry standardized signal playback apparatus not incorporating thedecoding features of the present invention. In addition, signals lackingthe encoding process features of the invention are likewise compatiblewith playback decoders which do embody the invention, and are providedsome overall enhancement.

Basically, the present invention is directed to various aspects of animproved encode/decode system for providing a predetermined balance orinterplay of gain structures, filter characteristics, various slew ratemodifications, and wave synthesis operations to reduce signaldistortions and improve apparent resolution. During the encodingprocess, an analysis of the signal to be encoded is made over time andthe results of this analysis are subsequently utilized in the encodingand decoding process to more accurately reconstruct the originalwaveform upon playback. This is accomplished while minimizing thedeleterious effects normally encountered in sampling and convertinganalog signals to digital signals and subsequently reconverting thedigital signals back to an accurate simulation of the original analogwaveform.

In accordance with the invention, control information developed duringthe aforedescribed waveform analysis is concealed within a standarddigital code and this information is subsequently used to dynamicallychange and control the reproduction process for best performance. Theseconcealed control codes trigger appropriate decoding signalreconstruction compensation complementing the encoding process selectedas a result of the aforementioned signal analysis. Since the controlcode is silent and the overall digital information rate is normallyfixed, the process can operate compatibly with existing equipment andindustry standards. In addition, and as previously indicated, signalslacking the encoding process features of the invention are likewisecompatible with playback decoders which do embody the invention, toprovide some beneficial enhancement.

To achieve higher performance with a fixed information rate, an on-goingtrade-off is made between dynamic range, to achieve improved smallsignal resolution, and peak level and/or slew rate, to achieve fastsignal response accuracy. These small change and fast change aspects ofa signal, as well as large and small amplitude aspects, each have theirown digital distortion or system compromise mechanisms. Since both largeand small aspects will not occur at the same time, an optimum encodingprocess or mix of processes favoring each signal condition can be chosendynamically, in accordance with the invention, to achieve an improvedsignal reproduction within a fixed digital information rate. A silent orhidden control code documents these changes from time to time in thesignal encoding process and is used to create the complementary level,slew rate, filter character, and waveform synthesis necessary to restorethe original signal during the decoding process.

In a presently preferred embodiment of the invention, the encoder systemhas much higher resolution and speed than the industry standard orencoded product, and is set up as an acquisition system with sufficientlook forward and look behind memory to compute the optimum processing ofthe signal and its corresponding reconstruction control code. Aspreviously noted, the processing of the signal is determined based on aconsideration of which trade-offs of resolution, speed, and level aremost appropriate for the signal conditions over time and how thereproducer can best be programmed to allow the most accuratereproduction of the original analog signal.

To be inaudible, the computed reconstruction control signal is encodedor encrypted to a random number sequence which may be insertedcontinuously or dynamically when needed into the least significantdigital bit or bits. The processed audio or signal becomes encoded tothe remaining bits.

Conventional decoding by a simple digital to analog converter of allbits of a recording encoded in accordance with the invention, yields asignal with slightly less dynamic range and only slightly higherbackground noise. However, the signal will have lower quantization andslew induced distortions and, hence, the processed encoded product, whenreproduced on non-decoding standard equipment, will sound equal to orbetter than an unencoded product.

A fully decoding player, in accordance with the invention, retrieves thecontrol signal and uses it to set up, operate and dynamically change acomplementary process to recover the pre-computed high accuracyinformation and provide low distortion reproduction of the originalanalog signal. Operations to do this include fast peak expansion,averaged low level gain reductions, selecting complementaryinterpolation filters, waveform synthesis, and others. When these areselected according to ongoing trade-offs, optimum for a particular setof signal conditions, an apparent increase of bandwidth and resolutionoccurs.

An improved digital system, in accordance with the invention, usesgroups of dynamically changing pre-determined performance trade-offsmade when signal conditions of the recorded program would createdistortion. Since digital distortions occur at extremes of high level,slew rate, and high frequencies, on one hand, and with quiet signals andshort small transients on the other hand, a best encode/decode strategyis chosen for that extreme without the process compromise hurting theopposite aspects of the program. To achieve this, the program is delayedlong enough so that a most likely distortion mechanism is identifiedprior to its emergence from the time delay, thereby allowing a bestencoding strategy and complementary decoding method to be determined andencoded. Performance is improved because any distortion compromise madeoccurs for opposite signal conditions, which are essentially nonexistentat that time.

In the simplest form of the system, an encoded dynamic range compressionand complementary reproduce expansion will improve performance.Furthermore, improvements are had by using averaged levels of smallsignals independent of their lower frequency and near supersonicfrequency spectral components to control processing providing improvedcomplementary restored resolution. In a similar manner, the strongestsignals receive processing having DC to maximum bandwidth forinstantaneous peak conditions, which also yields best complementaryrestoration. Only one correction need operate at a time and, hence,digital information is saved, or conversely, more apparent performanceis obtained from an unchanged digital information rate.

In addition, a further reduction of known and predictable digitaldistortions occurs by selecting a best low pass filter with the leastcompromise for program conditions during encoding and using acomplementary interpolation or low pass filter during reproduction.Also, other improvements are had from the reduction of known recurrentdistortions, such as transient errors, by synthesizing these componentsfrom lookup table curves of these distortions or missing information,and then scaling these to the signal at hand.

All of the aforedescribed improvements can also operate with varyingdegrees of success in a default or "open loop" mode at the reproducer bydetecting information about the encoded signal and then varying theseprocesses from the detected signal.

Digital systems typically have a very high signal to noise ratio, buthave a restricted working dynamic range of levels and restrictedfrequency response. The improved system of the present invention reducesdistortions and, as such, uses signal character dependent gain changes,filter optimization, slew rate processing, and waveform reconstructionor synthesis to do this. The improved system computes, within memory andprocess time limits, a continuously changing best compromise strategy ofavailable processes to give the best signal reconstruction. Thisobviously complex task yields a restoration control signal silentlyencrypted or noise disguised in a least significant bit code. Bycomparison, the reproducer system is simple, since its decoding andcomplementary signal restoration can occur with conventional multiplyingconverters, digital signal processors and other analog and digitaldevices similar to or already used in consumer electronics.

A conventional recording and reproducing digital system appearsrelatively simple and potentially accurate for all the data bitsencoded. In practice, however, using a very near to theoretical minimumsampling rate and the least acceptable number of data bits substantiallyaggravates speed and accuracy limitations from even the beststate-of-the-art circuits and components. In this regard, the worstoffenders are items such as filters, sample and hold circuits,analog-to-digital converters, digital-to-analog converters, and systemgrounding, timing and various process interactions and crosstalk.

The aforedescribed practical technological difficulties and theirpotential distortions can be greatly minimized by using higher samplingrates and more data bits than current standards allow. In fact, currenttechnological capability permits the reduction of cross-talk, timejitter and other noise interaction problems which, along with digitalbandwidth limitations, prevented the practical implementation of higherdata rates when current digital standards were first envisioned andestablished. With today's high speed converters operating much fasterwith more data bits, filters can become less severe and the greaterdifference between highest audio frequencies and the digital samplingrate then reduces beats, sideband foldovers, aliasing, as well as lossof small signal information. The present invention uses thesecapabilities by employing a high speed conversion process. The digitalinformation rate, though now much higher, can be computed, as an ongoingacquisition process, to an "error free" mathematically filtered lowersampling rate 16 bit code compatible with current standards. Mostdecimation oversampling encoders work like this. However, in addition,the invention anticipates alias, aperture, interpolation and amplituderesolution distortions from an "ideal" standard reproducer and computesthem during the encoding process for correction during reproduction.When the full process of the invention is used, even certain frequenciesabove the audio range or Nyquist limit of industry standard equipmentcan be sent through the system without creating sub-harmonic or foldoverdistortions. Hence, a closer to perfect record/playback system isprovided with minimal problems from filters, converters, and othercomponents or subsystems while remaining compatible with industrystandards.

For a Compact Disc system, "perfect" reproduction to 16 bit industrystandards will have a maximum of 65,536 well defined equally spacedresolution steps, each about 150 microvolts in amplitude when scaled tonormal professional audio levels (10 volts peak-to-peak maximum). Thisnumber, when stepped consecutively at the industry standard 44.1 kHzsampling rate, provides a slew of less than 7 volts per second. Fasterrates will skip numbers until, for a 10 kHz triangle segment, only 2.2sample points remain to define that waveshape as it would be filtered toits 20 kHz bandwidth. In this regard, more than a 1 giga Hertz samplerate would be required to include all 65,536 resolution points to createthat wave segment. Fortunately, an ideal interpolation filter will fillin all of these points provided the 2.2 samples have been timedaccurately enough. To do this to achieve a one half bit RMS averagedaccurate sample of a fast changing signal the sample timing must occurwithin: ##EQU1##

This sample, accurate in time and amplitude, must be held long enoughfor conversion to digital code. Usually, a charge on a capacitorrepresents this information. However, most dielectrics and insulatorsused to fabricate capacitors have complex losses as well as past historymemory which create a complex delayed voltage change, fieldre-distribution errors and leakage. When abrupt changes in level fromsample to sample occur, as they do with sampled high frequency audiosignals, these errors often become much greater than when signal levelsdon't change. To have less than a half LSB of RMS averaged error thehold accuracy becomes: ##EQU2## or about 2.3 u Volt per u sec.

Such performance is well beyond simple applications of most modernelectrical passive components, much less integrated circuits. Obviously,practical consumer playback equipment will not do better, and theresulting errors can produce slew rate related transient intermodulationdistortion components, which are among the most audibly objectionable.Specifically, these result from acquisition time uncertainty or jitter,slew rate related non-linear switching offsets, various types ofdielectric hysteresis causing previous event related errors, polaritydependent sample discrepancies, and unpredictable hysteresis withinconverters as well as other factors. Thus, practical systems often havecomplex signal related errors as high as twenty times more thantheoretical resolution limits of the current 16 bit standard. Hence, aprocess providing more sample points per second with the least voltagechange per sample will yield a signal with lower transientintermodulation distortion.

A second distortion mechanism occurs with very small signal amplitudechanges of about 5 to 20 millivolts represented by digital activities ofless than about 8 bits in a typical 16 bit system. These levels seldomoccur by themselves yet can still be a small but audible part of alarger low frequency dominated signal. Hence, these small signals canoccur averaged at many different voltage levels or digital numbers of alarger slow waveform. A practical example of this would be midband hallreverberation decay and bass sounds combined. The reverberation signalattenuates and sometimes completely disappears as it becomes chopped orbroken segment parts of the bass waveform. As previously indicated,these breaks represent the 150 uV resolution limits of a "perfect" 16bit reproducer. In practice, very small signal changes can becomestepped outputs, or more often distort to irregular step to step changeswith an uncertainty or hysteresis which occurs due to errors withinconverters and from external interference and crosstalk. This produces acollapse of the sense of space in a recording and generates impulsivegrainy noise effects which are usually made less objectionable by addinga random noise voltage to the signal prior to encoding so that the steperrors become randomized from the uncertain samples created. Thus, thestepped or quantized distortion becomes a less objectionable noisemodulation and the least bit signal cut-off levels are now smoothed to agradual gain loss with progressively smaller signal changes. A betterform of distortion reduction occurs by increasing the sample points perunit voltage change. Unfortunately, like the process to increase slewaccuracy, a much higher digital information rate than that of thecurrent standard is needed to accomplish this.

Low signal level digital errors produce distortions such as quantizationnoise and resolution loss. Whereas, high signal level high frequency andslew rate related errors produce distortions such as sporadic beats andfast signal change envelope related subharmonics, referred to astransient intermodulation distortion or TIM. One is easily misled bytest signals with a continuous envelope nature, in that they tend toaverage over time and cancel many of these distortions and thereforeincorrectly indicate only very small resolution and converter inaccuracydistortions. Unfortunately, waveforms like those in music continuallychange and, as noted, may provide much higher and far more objectionablenon-harmonic TIM and resolution problems.

Digital distortions occur with high slew rate and small amplitude signalchange conditions and, as previously indicated, both are not likely tooccur at the same time. Hence, in accordance with the invention, thesystem identifies either a fast slew or a small change character of thesignal waveform and implements the appropriate corrective process.During encoding, the nature of program signal changes can then determinewhich corrective process is used as well as a best reproduce conjugateor process at any time during decoding. One process can borrowinformation rate from a less needed performance capability whenpotentially severe distortion conditions in the signal call for it. Inthis manner, a decision to provide more points per fast voltage changeyields an equivalent higher sampling rate at the expense of lessimportant low level resolution. Conversely, a smaller voltage change persample automatically reduces the momentarily unneeded speed capability.Such interplay and compromise can be managed and/or computed to maintaina substantially constant digital information rate. Under thesecircumstances, the processed, decoded, analog output may have anapparent increase of bandwidth and resolution and, as noted earlier,when these improvements occur, one or the other as needed, thefundamental causes of digital distortions as well as their effect onimperfect reproducers can be reduced.

A similar correction strategy is applied to reduce filter trade-offcompromise errors between transient response, phase accuracy, settlingtime, group delay, and other distortions inherent with filteringmethods. Such errors may not be non-linear, and hence, will not appearas harmonic distortion; however human hearing is sensitive tomanipulations of waveform shape and to the settling time of complexsignals. Typically, the smallest amplitude high frequency signals arelikely to have excessive transient ringing and process noises fromaggressive filtering, whereas sub-harmonic beats and other filteringnoises may occur with intense high frequency signals. The instantaneousversus non-instantaneous character of complex signals is reproduceddifferently from one filter type to another. As before, the same largesignal/small signal selection criteria hold, allowing a best encode anddecode filter choice, without having to compromise for the opposite,essentially non-coexistent, program conditions.

Hence, the method and apparatus of the present invention utilize apre-calculated optimal interplay of gain, slew, filter selection, andwaveform synthesis operations done individually or as a composite allinclusive process which becomes encoded and decoded in a complementarymanner to reduce distortions and improve resolution. Included in such asystem is a record compress--play expand system with some featuressimilar in ways to those used in noise reduction systems. Most suchnoise reduction systems use either peak or RMS detectors to examine theincoming signal and convert its level to either fast or slowly changinginternal DC control signals which ultimately drive a transient freeswitching element or an analog variable gain device. When set up forgain reduction, with increased input signal level, the output signal iscompressed so that tiny signals are amplified and strong distortionprone signals are attenuated. Upon playback or decoding, a similarcircuit set up for gain expansion, detects level changes and restoresthe signal to an approximation of its original dynamics.

In contrast to traditional noise reduction, the system of the presentinvention corrects distortion. It does this by altering gain structure,as well as amplitude and slew rate linearity, for extreme low and highlevel signal conditions. Low level, small changing parts of the signalare detected and used to control the gain of the whole signal which thenincludes more encoded bits. This gain control is derived from a broadmiddle spectrum of the signal and is active at signal levelsrepresenting the lowest levels perceived by human hearing. It is notactivated by low frequencies, near supersonic frequencies, or whenhigher level mid-band signals are present. In this manner, the gainstructure increase maintains a minimum LSB dither-like activityindependent of inaudible sounds and maintains ambient and backgroundinformation as well as masking quantization and monotonistic errordistortions previously described.

Infrequent peak levels are instantly compressed with a transfer functionhaving very low distortion for signals near maximum level and producingminimum upper harmonics once the limit threshold is traversed. This typeof operation does create an occasional higher distortion on peaks,however it prevents catastrophic overloading during recording and allowsa higher recording level with overall lower distortion.

Infrequent fast slew portions of the waveform can be expandedsymmetrically in time, and/or in samples, to encompass more encodedbits, and, as before, other parts of the waveform may be unaltered. Thisoperation may be a dispersion process where time delay is altered, or itcan be a graphical waveform synthesis. It takes an instantaneous eventand spreads it in time, and like the peak limiter, it creates distortionin undecoded playback.

Gain change, peak limit, and slew rate compression operations and theircomplements or restorative operations are practical with analog ordigital techniques. Voltage controlled amplifiers, diodes, delay lines,and chirp filters, and multipliers are typical analog building blockswhich can be assembled to create these functions. Equivalent digitalsub-routines and dedicated process algorithms and components are alsoavailable. Distortion free digital processing is complex; for example,rounding off errors may have to be dithered and interpolated over time.However, once implemented, digital operations are very stable andprecise compared to the variables subject to tolerances and adjustmentsrequired for the analog control of gain, dispersion, bandwidth and timeconstants.

The aforedescribed level and slew processes of the present inventioncorrect distortions occurring from opposite signal conditions which arenot likely to occur at the same time. Hence, these can interplay and atmaximum correction capacity can borrow from an opposite less neededperformance capability to maintain constant digital information rates.The wave synthesis process of the present invention operates with knowndistortion waveshapes which, when encountered during encoding, aresubsequently called out of memory by code for complementary correctionduring reproduction.

Level and slew correction works for known signal conditions havingunpredictable distortions and synthesis works for known distortionsoccurring from signal conditions unpredictable at the reproducer. Unlikeeven state-of-the-art noise reduction processes, this system'sprocessing is under intelligent control and given sufficientcomputation, trial and error, or successive approximation time, the bestcorrection scheme and its encoding for reproducer process control isreadily determined and optimized.

Wave synthesis, in accordance with the invention, is a keyed operationused to recall from memory a number of predictable and/or recurrentdistortions known to occur at the reproducer. Small waveform segmentsfalling outside of the Nyquist sampling limits, repeated quantizationdistortions, and interpolation filter parameters can be recalled from alook-up table in memory or synthesized from information sent in thehidden code, and used for improved playback. The synthesis memory cancarry several interpolation waveshapes which best connect points at andbetween samples. These larger waveforms will maintain theircharacteristic shape independent of level, just as the reproduced signalwould do. Once the connecting waveshape has been recalled from ROM, itmust be scaled to fit the signal. Since only very slowly changingwaveforms will have samples without bit resolution levels in between, aform of level detection is necessary to make synthesized segments scaledto the signal. What would have been level detectors and gain controlleddevices in an analog system are replaced by equivalent digital signalprocessing functions in a digital system. Once this has beenaccomplished, the reconstructed waveform has more equivalent data pointsin time and level and, when pre-computed properly, a lower distortionresults from the curve fitting.

In light of the foregoing, a practical system, in accordance with theinvention, may have many times better signal resolution and much betterfast transient signal accuracy. A much greater digital information ratewould normally be necessary to achieve these results. Data is saved byprocessing only distortion producing conditions. As noted, resolution isselectively and adaptively traded off for slew accuracy and slew rate ormaximum level is borrowed for higher resolution. Information rate isconserved by toggling back and forth or fading from process to processwhen needed.

It should also be apparent that implementation of various subsystemdesigns may be in either analog or digital form, monitoring and analysisof the waveform may be accomplished at varying locations in the systemincluding the reproducer and in either analog or digital form, otherparameters of the waveforms may be selected for compensation, andcontrol codes or other waveform corrective message information may beinserted and extracted in a variety of different ways, without departingfrom the basic concepts of the present invention.

Hence, the method and apparatus of the present invention forencoding/decoding signals with minimal distortion satisfies a longlasting need for a compatible system which provides an adaptiveinterplay of gain, slew rate, filter action and wave synthesis processesto substantially reduce signal distortions and improve apparentresolution.

The above and other objects and advantages of the invention will becomeapparent from the following more detailed description, when taken inconjunction with the accompanying drawings of illustrative embodiments.

DESCRIPTION OF THE DRAWINGS

FIG. 1 is an overall block diagram of an analog to digital encodingsystem in accordance with the invention;

FIG. 2 is an overall block diagram of a digital to analog decoding andreproducing system in accordance with the invention;

FIG. 3 is an more detailed block diagram of an example of an analog todigital encoding system in accordance with the invention;

FIG. 4 is an more detailed block diagram of an example of a digital toanalog decoding and reproducing system in accordance with the invention;

FIGS. 5a through 5e graphically depict waveforms illustrating samplingand encoding errors encountered with low level and rapidly changingwaveforms;

FIGS. 6a through 6f graphically depict various signal waveforms duringthe limiting and reconstruction of a triangle wave, in one embodiment ofthe invention;

FIGS. 7a through 7d graphically depict waveforms illustrating varioustypes of distortion encountered with different types of filters;

FIG. 8 is a block diagram of a processing system in accordance with theinvention, using analog processing technology;

FIG. 9 is a block diagram illustrating filter selection control in oneembodiment of the invention;

FIG. 10 is a block diagram of a process switcher utilized in oneembodiment of the invention;

FIGS. 11 and 12 graphically depict waveforms and distortion plotsillustrating system response before and after process for two filtertypes, in accordance with the invention;

FIG. 13 is a block diagram of an analog implementation of a slew ratecompression and expansion system;

FIGS. 14a through 14e show waveforms illustrating the operation of aslew rate compression and expansion system;

FIG. 15 is a block diagram of a more advanced, presently preferreddigital embodiment of the encode system, in accordance with theinvention; and

FIG. 16 is a block diagram of a more advanced, presently preferreddigital embodiment of the decode system, in accordance with theinvention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention is directed to a system for an electronic methodand apparatus for signal encoding and decoding to provide ultra lowdistortion reproduction of analog signals, while remaining compatiblewith industry standardized signal playback apparatus not necessarilyincorporating the decoding features of the invention. The improvedsystem provides a selective interplay of gain, filter selection, slewrate and wave synthesis operations to reduce signal distortions andimprove apparent resolution from a recorded product, under the controlof concealed or silent control codes when necessary for triggeringappropriate decoding signal reconstruction compensation based upon aprevious signal waveform analysis made during the encoding process forthe recorded product. In addition, signals lacking the encoding processfeatures of the invention are likewise compatible with playback decoderswhich do embody the invention, and are provided the benefits of someoverall enhancement based upon a signal waveform analysis made duringplayback.

Referring now to the drawings, and more particularly to FIG. 1 thereof,there is shown, in general terms, the analog to digital conversion andencoding subsystem of a typical recording system embodying features ofthe present invention.

As shown in FIG. 1, an analog signal 99 is directed as input to aprocessing subsystem 100 which converts the analog signal into digitalform, including such tasks as filtering, sample and hold, analog todigital conversion and the like. The digital output 100a from thesubsystem 100 is directed to two subsystems, a memory subsystem 101 andan analysis and computation subsystem 102. In memory subsystem 101, thedigital signal is delayed or stored for further use and manipulation.The digital signal output of memory subsystem 101 is sent to subsystem102 at input 102E. Using the output of subsystem 100 at input 102A, thewaveform analysis and correction computation subsystem 102 continuouslymonitors and evaluates the digital format waveform as it is being storedin the memory subsystem 101 in order to determine the physicalcharacteristics of the stored waveform ultimately to be reconstructedand the required corrections necessary for accurate reconstruction andrestoration of the original analog waveform 99. This evaluation relatesto reconstructive level, slew, and waveform synthesis requirementsultimately to be provided by complementary compensation in anappropriate decoding and signal reproduction system (FIG. 2). Theevaluation may also predict alias components for subsequent conjugateneutralization. Some aspects of the signal evaluation may be performedon the analog signal by subsystem 100, and the results sent to subsystem102 at input 102d.

The corrective procedures are applied to the digital signal from thememory subsystem 101 by subsystem 102 under the control of signalsresulting from the analysis. The process controller 102 also generatescontrol codes for use by the decoder which are converted to properformat and appropriately encrypted into the digital signal so that thecontrol codes can silently ride along with the digital representation ofthe original analog waveform 99 and be provided as an encoded digitaloutput 103. Some of these corrective procedures will relate, not just todistortion characteristics occurring as a result of the basic conversionof the analog waveform itself, but also to procedures deliberatelyintroduced by the encoder for subsequent complementary decoding, such aspeak limit/subsequently expand for high level signals and averagedcompress/subsequently expand for low level signals.

As best observed in FIG. 2, there is shown, again in general terms toillustrate some of the basic overall concepts embodied in the presentinvention, a digital to analog conversion and decoding subsystem of atypical reproducing system embodying various features of the presentinvention for reconstructing the original analog waveform.

In FIG. 2, the encoded digital signal 103, recaptured from anyappropriate recording medium (not shown) such as tape or disc, isdirected as input to a digital signal analysis and processing subsystem104 and to memory subsystem 107, which delays the digital signal. Signalanalysis subsystem 104 extracts control code information inserted in thesignal at the encoder and may also analyze the signal itself todetermine its characteristics. These operations include appropriatemeans for control code detection, signal filtering, level detection,spectral analysis and the like. The detected control codes and signalanalysis in the processing subsystem 104 are used to generate controlsignals directed to a reconstruction compensation subsystem 105 whichinteracts with the processing subsystem 104 and operates on the delayeddigital input signal 108. Subsystem 105 includes digital to analogconversion, and may include further memory, such as one or more ROM's orlook-up tables, for various types of reconstruction compensation used,in accordance with the invention, to correct the digital signal 103.

The compensation subsystem 105 typically will respond to the variouscontrol codes, or the absence thereof, to generate a variety ofcorrective compensations such as slew rate, level, filter selection, andwaveform synthesis which, through appropriate interaction with theprocessing subsystem 104, yields a reconstructed analog signal 106 withminimal distortion and enhanced apparent resolution, all without theneed for increasing industry standardized digital bandwidth.

It will be appreciated by those of ordinary skill in the art that thesystems of FIGS. 1 and 2 are merely illustrative of simplified generalapproaches for practicing certain basic aspects of the presentinvention, and implementation of the systems of FIGS. 1 and 2 may take awide variety of specific forms without in any way departing from thespirit and scope of the invention.

It should also be apparent that implementation of various subsystemdesigns may be in either analog or digital form, monitoring and analysisof the waveform may be accomplished at varying locations in the systemand in either analog or digital form, other parameters of the waveformsmay be selected for compensation, and control codes or other waveformcorrective message information may be inserted and extracted in avariety of different ways, without departing from the basic concepts ofthe present invention.

By way of example, one possible implementation of the general structureabove is presented in more detail in FIGS. 3 and 4. These drawingscorrespond to FIGS. 1 and 2, and illustrate more internal detail.

Referring now more specifically to FIG. 3 of the drawings, there isshown an analog to digital encoding system in accordance with theinvention. Analog input signal 99 is applied to a buffer amplifier, thefirst element of the analog to digital subsystem 100. The output of thebuffer amplifier drives an analog low pass anti-alias filter, whichremoves any high frequency components of the input signal falling abovethe Nyquist limit of half the sampling frequency. The output of the lowpass filter has an analog dither signal added to it and then it isapplied to the input of a sampling analog to digital converter. In theconverter, the signal amplitude is sampled at regular intervals and theamplitude of each sample is converted into a number or digital word. Theseries of digital words from the converter make up the digital signal,which is sent to the analog to digital conversion process controller.This process controller has generated the dither signal which was addedto the analog signal before conversion, and, typically, the controllersubtracts the dither from the digital signal, giving a vernierenhancement to the conversion accuracy as well as spreading anyconverter nonlinearities into a noise-like signal. The ADC processcontroller may also make other corrections or additions to theconversion process, such as noise shaping. The output of this module isa high resolution digital signal 100a which is sent to subsystems 101and 102. It should be noted that this digital signal has both higheramplitude resolution and greater sampling rate or time domain resolutionthan the industry standard digital signal which is the final output ofthe encoding system.

Memory subsystem 101 is used to delay the high resolution digital signal100a before sending it to 102e. This time delay gives subsystem 102 timeto analyze the signal and choose appropriate corrective procedures to beapplied during encoding.

The high resolution digital signal from subsystem 100 is also sent tothe signal analysis process controller unit of subsystem 102 at input102a. This unit analyzes the characteristics of the signal as it isbeing stored in the delay memory 101 and makes decisions about employingcorrective procedures such as instantaneous peak amplitude limiting, lowlevel gain compression, choice of best "brick wall" low pass filter,transient reconstruction and so forth. The unit then sends commands 102bto the units which process the delayed digital signal to carry out thecorrective procedures. The signal analysis process controller alsogenerates a control code 102c which it sends to the code encryption unitfor addition to the output signal. This control code tells the decodesystem what has been done and how to recover an accurate representationof the original input signal.

The delayed high resolution digital signal from the memory subsystem 101is sent to the decimation filter unit at 102e. Here, the oversampledinput signal is decimated down to the industry standard sampling rate.The choice of optimal filter characteristics is dependent on the natureof the program signal at the time. Such factors as transient content ofthe signal, presence of large amounts of alias producing highfrequencies, etc. are taken into account by the signal analysis processcontroller, and a filter control signal 102b tells the decimation filterwhich parameters to use. The output of the decimation filter has theindustry standard sampling rate and very high amplitude resolution. Itis sent to the level control processing unit.

The level control processing unit uses such operations as instantaneouspeak level compression and low level average gain compression to squeezethe high amplitude resolution of the signal into the industry standardresolution (such as 16 bits for CD). These operations are done under thecontrol of the signal analysis process controller. The level controlunit may also include other techniques such as the addition of digitaldither to allow resolution below the least significant bit level andtransient time domain or slew rate compression. The output of this unitis sent to the silent code encryption unit.

The silent code encryption unit takes the control codes 102c from thesignal analysis process controller, which are commands and informationfor the decoder system, and adds them to the digital signal. One methodof doing this involves encrypting them into a pseudo-random noise-likesignal and inserting it as needed into the least significant bit of thedigital signal. Other methods include the use of "user" bits in standardcode or unused bit combinations which may appear to be errors to anormal decoder. The common characteristic of these methods is that theyprovide a silent side channel for control information which rides alongwith the program digital signal.

The final task of the code encryption unit is to encode the compositedigital signal into an industry standard format for recording, etc. Theoutput of this unit is a standard digital signal 103, which, forinstance, could be sent to a recorder. This completes the description ofthe encoding system.

Referring now to FIG. 4 of the drawings, there is shown an example of adigital to analog decode/reproduce system in accordance with theinvention. The input digital signal 103, from a tape recorder, CD, etc.,is applied to the signal and code analysis subsystem 104 and to memorysubsystem 107.

Memory subsystem 107 provides a time delay for the digital input signalin order to allow subsystem 104 time to do its analysis. The delayeddigital signal output 108 of the memory subsystem is sent to the levelcontrol unit of subsystem 105.

The digital input signal 103 is also applied to the signal analysis,code analysis and process control subsystem 104. This subsystemseparates from the signal the silent control code inserted by theencoder. This control code contains information about what processingchoices were made by the encoder and what complimentary correctionsshould be applied to reconstruct the most accurate reproduction of theoriginal analog input signal. The subsystem may also analyze the signalitself to determine the best reconstruction strategy, measuring suchparameters as the signal amplitude, spectral content, etc. The subsystemthen generates a series of control signals to control the various unitswithin the reconstruction processor 105, each of which performs aspecific type of operation on the program signal.

The reconstruction compensation and digital to analog conversionsubsystem 105 is made up of a number of processing units which operateon the program signal under the control of the analysis and processcontrol subsystem 104. Each of these units has a control connection to104. The first of these is the level control unit, which receives thedelayed digital input signal 108. The level control unit performsamplitude scaling complementary to that performed by the encoder, suchas instantaneous peak expansion and signal averaging based low levelexpansion. The output of this unit is a digital signal at the inputsampling rate, but with higher amplitude resolution than the input. Thisoutput is sent to the interpolation filter unit.

The interpolation filter unit creates an oversampled digital signal byinterpolating between the points represented by the input signal. Thebest filter parameters for this interpolation are chosen dynamicallybased on the control codes, and possibly also signal analysis, so thatthey complement the parameters of the decimation filter in the encoder.Other processing such as noise shaping and transient reconstruction mayalso be done by this unit. The output signal of this unit is a highresolution oversampled digital signal which is sent to the digital toanalog converter unit.

The digital to analog converter (DAC) unit converts the high resolutiondigital signal into an analog signal. It may be a standard converter ora multiplying converter which is used to further effect level changes inthe signal. The output of this unit is an analog signal which is sent tothe analog processing unit.

The analog processing unit contains an analog interpolation filter andbuffer amplifier. It may also contain other processing, such as levelcontrol, under the control of the analysis and process control unit.Since it is operating on an analog signal, the control signal may beconverted to analog form in the control DAC before being applied here.

The output of the analog processing unit is an extended range analogsignal 106, which is a close replica of the original analog input signal99. The overall system of the invention makes possible a more accuratereconstruction of the original analog signal than would have beenpossible from conventional systems using the same digital recordingstandards.

The ensuing analyses and discussion are intended to provide furtherbackground for a proper understanding of the practice of the inventionand to further illustrate and describe a variety of analog/digital modespresently contemplated as feasible for carrying out the invention.

Analog to digital encoding, in accordance with one aspect of the presentinvention, works as a sample rate down converter which allows resolutionenhancement and reduced filter artifacts. Analog to digital conversionis made at a high sample rate and extended bit resolution, both wellabove that required for the final product encoded format. This highdensity code is then arithmetically processed back to the desired enduse number of bits and sample rate. With this arrangement, manyadvantages occur. Analog "brick wall" low pass filtering is unnecessaryas the very high ingoing sample rate allows much more gentle andphase--time domain controlled audio band cut off before Nyquistdistortions occur. The sample and hold--analog to digital convertersubassembly can be supersonically dithered in a known and controlled wayto create duty cycle modulated low level code parts in which anymonotonicity or missing code errors are spread out as noise sidebandsaround the dither signal. These will be very high frequency and, hence,almost inaudible, unlike signal related noises of standard systems.

An important advantage is that the "brick wall" low-pass filter requiredto prevent Nyquist--Alias errors can be implemented as a digital filter,which has highly reproducible characteristics free from phasedistortions. The characteristics of this filter can be chosendynamically based upon an analysis of the high resolution signal tominimize distortion. Hence major filter and analog to digital encodesystem problems such as pre-echo, transient ringing, group delayanomalies, missing code errors, alias distortion and beats are greatlyreduced or eliminated.

A very powerful arithmetic "engine" and operating program, as well asdigitally operated feedback and feedforward parts are utilized to makethe wideband audio to selected format digital conversion. However, sucha system also easily performs instantaneous high level limiting andaveraged low level expand operations. In practice, approximately anextra 4 bits dynamic range can be had from systems having complementaryplayback and significantly improved sonics will occur from standardunprocessed compatible playback systems. This is because the recordingengineer can raise levels without overload problems, thereby simplifyingrecording sessions and very low level ambient information will alwaysmaintain least bit activity to prevent monotonistic errors. Bothprocesses on a "perfect" system would be inaudible. However, on realdigital systems the sonics will improve, as the slightly higher levelsreproduce with lower distortions through digital systems. Both theinstantaneous peak limit/expand and the averaged compand/decompandfunctions are controlled so that the degree of processing can becomputed and automatically controlled as needed for best programreproduction. This configuration allows very fast corrective action tobe varied by a low bandwidth control signal. This control signal can behidden within error correction codes, placed on other audio channelswithin the system, or random noise encrypted and inserted as needed intothe least significant bit or bits. In this regard, the benefits farexceed any added error which is below the practical resolution limits ofmost equipment.

Basically, one aspect of the system of the present invention addressesand partially corrects several distortions known to occur with A to Dand D to A conversions of complex signals. Some of these errors arehardware related and are correctable with more exacting methodology.Other distortions are the result of the bit depth and sampling ratefixed by industry standards and are minimized creatively by the system,by varying dynamic optimization between performance aspects. Determiningthe best form of optimization can be very complex, as many suchdistortions do not occur with the steady state type signals used fordistortion tests, and must be minimized to subjective criteria. Most aretransient intermodulation distortions (TIM) of which certain types areobjectionable 50 to 60 dB below the program material. As will be shown,the hardware mechanisms for producing these distortions are non-linearswitching and hysteresis in capacitors in sample and hold circuits,digital to analog crosstalk, slew rate asymmetry from large numbers ofparts in the signal path and crosstalk between analog and digitalsignals.

The encoding process of the invention makes program signal alterationswith the least audible consequence for reproduction from standardequipment. These changes reduce certain types of distortions andincrease signal resolution, thus providing improved playback, e.g.,better spatial sense and less brittleness. Using the decoding process ofthe invention, the reproducer can be made to track and inverselycompensate these signal alterations thereby allowing substantially"exact" playback, with greatly reduced distortion.

In order to provide a further understanding of the problems with digitalsystems and how they are resolved by practice of the invention, thedifficult technology areas and distortion mechanisms occurring fromideal as well as practical implementations are presented as follows:

1. Resolution limitations with small signals:

Distortions in digital systems increase with decreasing signal levels,and the smallest signals become broken and tend to disappear. A goodanalog system and a 16 bit digital system can both handle signals withan 85 dB dynamic range. Typical analog systems have infrequent highdistortion at signal peaks whereas digital systems have continuousdistortions at low levels. Experience has shown these distortions to beaudible, hence some form of noise dithering is often used to smooth outquantization steps and allow information fill-in. This technique createsnew distortion from beats between dither, sampling and signal frequencydifferences. Nearly inaudible low level upper harmonics then create moreperceptible low level sub-harmonic interferences.

FIGS. 5a,b,d Sampled, encoded, and decoded low level low frequencysignal.

FIG. 5e Above dithered with high pass filtered random noise.

A unique solution to the aforedescribed problem is provided by thesystem of the present invention.

In this regard, a minimum low level signal activity is maintained at alltimes by using a gain expanding device or its equivalent digital processto increase the gain of the system only when the average level of thesignal is low. This low level gain riding is programmed to maintain aminimum LSB dither-like activity which will tend to mask quantizationeffects or other least significant bit monotonistic discontinuities. Thereproduced program will sound better on less expensive players notincorporating the invention, which often have high distortions fromthese kinds of problems. For exact decoding, the invention provides anopposite gain-structured average level compression device or equivalentdigital process which restores the full dynamics of the original signal.When this complementary process is used at the reproducer, low levelgain reduction occurs and quantization noise is reduced.

Best operation of the process occurs when gain control is based upon abroad middle audio spectrum, as this restriction prevents gain pumpingfrom noise and bass fundamentals. An average level detector controls avariable gain function as follows: An RMS detector or other averagingtype device receives these signals. Gain boost is determined from theband restricted program and its average boost level is controlled byattack, sustain and decay parameters much like those in synthesizers. Toprevent overload from sudden signal changes, the unmodified fullbandwidth program is delayed long enough to allow time constants neededfor the averaging process to respond and anticipate events. The delayedprogram is then gain controlled either by adding the signal to itself orby multiplying. These methods provide the benefit of undecodedreproduction with the lease audible artifacts.

A decoding reproducer can operate in the aforedescribed manner exceptthat it performs a gain reduction in response to its own determinationof average signal level, using a control signal detected from a reducedbandwidth version of the input signal, having attack, sustain, and decaytime constant averaging to operate a gain control process passing thedelayed program. This default or non-controlled mode can be madesufficiently accurate since the recorder has the same building blocksand can test the reproducer response for a program event and make touchup corrections prior to the full band signal reaching the variable gaindevice.

This system operates like many compress/expand type systems. Variousinternal operations like filtering, detection, gain control, integrationfor time constants, time delays, log conversions, and curve generationcan be made operable from functional modules or from known digitalprocess programs. An analog system can be constructed from such buildingblock functional units. Each unit is fully buffered, gain structured,and instrumented allowing many different types of systems to be set upeasily. A more detailed description is found below as part of thecircuit description.

2. Catastrophic overload from large signals:

Large signals can suddenly overload a digital system to create severeunmusical crackling and other breakup sounds. Most analog recordersgradually overload with program related harmonics, are more forgiving,and therefore work closer to their maximum capability. Recordingengineers using digital equipment typically will choose moreconservative levels and risk the resulting lower resolution and higherdistortion.

FIGS. 6a-f Digital/Analog overload on triangle wave.

A unique solution to this problem is also provided by the system of thepresent invention.

In this regard, a peak limiter is used to imitate analog overload.Higher peak distortions occur infrequently, but now the average programlevel can be higher, and the average percent distortion is usuallylower. Exact program reproduction is restored by a peak expansion havinga conjugate transfer function to that used to limit the program. Thispeak limiting can be applied to the signal either in the analog domain,before it is converted to digital, or in the digital domain, and theconjugate expansion can likewise be applied in either domain. Thepreferred method is to do both operations in the digital domain, sincethe expansion can then be made to track the compression exactly, and theshape of the limiter curve can be controlled for least distortion onundecoded playback. In order to make the scheme work effectively in thedigital domain, however, one must have an analog-to-digital anddigital-to-analog converters with sufficient amplitude resolution tohandle the dynamic range.

If the peak limiting operation is applied to analog signals at theencoder input it can be digitally conjugated to yield a linear signal ofmore data bits. This effectively creates an A-to-D converter with higherresolution which has higher errors for signal peaks where the diodecompression occurs than for the majority of its dynamic range.

Building block or functional module circuits can be hooked up to performthe input and output analog operations. Voltage controlled amplifiers,four quadrant multipliers, log-exponential converters, and multiplyingDAC systems are available. Most of these devices make a more logical anddirect implementation, but they also have temperature tracking problemsand most of them create higher noise and distortions than the method weuse.

In the digital domain, DSP programs can work from lookup tables,arithmetic sub-routines, and process combinations. Just like low levelaveraging this operation can test decoder response, determine a fix, andthen make the correction to a delayed data signal. Certain types oftransient ringing occurring after decimation will create somere-construct distortion to very large signals. These errors are similarto those in other dynamic range enhancement systems. They appear to bean accepted compromise of dynamic range enhancement and for now we havenot dealt with the problem.

3. Limitations of low pass anti-alias filters:

The industry standard low sampling rates force a very narrow transitionregion between pass and reject responses of anti-alias filters. Thisvery greatly increases complexity of either analog or digital filterimplementations and prevents having all aspects of filter performanceoptimal in one design. A compromise results. Best resolution forextended high frequency response and Nyquist rejection necessitate poortransient and time settling response. Less aggressive filtering givesless high frequency extension with improved transient settling or givesimproved high frequencies and poor alias rejection. Filter sonics aredifferent from one to another and each works best with certain types ofprogram material. The filters shown below are symmetrical, fixed groupdelay types made from large numbers of sections representative of goodfinite impulse design practice.

FIG. 7b Low Alias filter: extended maximally flat high frequencyresponse, maximum alias rejection, steep transition region.

Electrical: Rapid changing high frequencies have unsymmetrical sidebandswhich create vestigial amplitude ripple effects. Long settling time fortransients.

Sonics: Inner detail sound is compromised to get low alias distortion athigh signal levels.

FIG. 7c Compromise filter: Reduced high frequency response, largertransition region, reasonable Nyquist rejection.

Electrical: slower transient response with faster settling, less flatfrequency response.

Sonics: Dull soft dynamics, moderate inner detail, moderately clean highfrequencies--best for simple low level signals.

FIG. 7d High resolution filter; good transients, but peaked highfrequencies and poor alias rejection.

Electrical: High frequency response has a dip and then a peak, producingsteep transients and fast transient settling time. Complementarydecoding yields very fast transition speed to infrequent transientedges.

Sonics: Best for complex signals when alias distortion does not createproblems. Poor cymbal sound.

Near ideal digital or analog implementation of the above filters istheoretically possible. Both are characterized by similar equations. Noone of the above works best for all situations; each has itscompromises.

The aforedescribed problem is likewise resolved by the system of thepresent invention.

Observe the program data and dynamically choose a best encoding filterautomatically, based on the program content. Send to the reproducercoded control signals indicating these filter choices. This makes itpossible for the reproducer to initiate its own complementary orconjugate response to the encoding filter characteristics.

Filter corrections require an operational system consisting of encodefilters, decode filters, selection logic, a means of switching from onefilter to another, time delays, and a means of encoding control signalsfor the reproducer. Each of these can be performed in the analog ordigital domain and quite often easy processes in one are quite difficultin the other. A brief description of each subsystem follows:

Digital Filters:

Complex filters are best created by specialized DSP chips. A typical DSPchip is configured with 24 bit multipliers and 56 bit accumulators in afunctional configuration which is very efficient at performing themultiply and add operations required for digital filter algorithms. DSPchips can be used to make symmetrical finite impulse response filters,convolution networks, spectrum analyzers etc. In most instances, afilter equation is designed using a computer to simulate its responseand the resulting coefficients are then incorporated into a program forinsertion into the DSP chip's memory. PROM chips are programmed withthese numbers and connected to the DSP, or for volume production anequivalent mask programmed ROM may be used, which may even be residentin the DSP chip. Hence, different filter responses can be achieved bychanging to different program coefficients or different programs, allstored in ROM, or by using several separate DSP processors, each runninga single filter program.

This approach applies to decimation, which reduces oversampled data toan "alias free" lower sampling rate and is used during encoding, or tointerpolation, which produces an oversampled signal and may be usedduring the decoding process. Both involve the use of low pass filtersand both rely on multiple or oversamples of a base or industry standardsample rate.

Decimation takes a stepped "curve" of multiple samples for each oneneeded and finds the best number for each sample at the desired rate anddiscards the rest. Since waveform reversals within each final sampleperiod do not reproduce and only create distortion because of theNyquist limit, a low pass filter operation must remove the aliasfrequencies of such events, eliminating the audio consequences. In onesystem, we use a 16 bit converter operating at an 8 times oversamplerate, and in another, an 18 bit converter operating at 4 timesoversample rate. In theory we gain 2-3 bits from subtractive dither withoversampling and gain another two from limiting. During any one sample,hundreds of DSP operations on these bits will have been in process andaccumulated to produce an encoded low pass filtered number 24 bits long.For our system, 20-22 of these bits can have useful information.

Interpolation takes each sample and creates a stepped "curve" of manycomputed intermediate values or oversamples. Here the intent is toreconstruct a waveform like the original signal. As before, hundreds ofDSP operations are needed and again fully processed points betweensamples as well as any computed signal restorations become added bits tothe DAC. With the system of the present invention, we then get 24 bitsat an 8 times oversample rate. Of these 24 bits, approximately 20 bitscontain useful information, and digital to analog converters with thisresolution are just becoming commercially available.

Because of DAC performance limitations, it may be necessary to handlethe added bits from limit restoration and low level averaging during orafter digital to analog conversion. In one version of the system, an 18bit 8× sampling DAC and noise shaping is used to achieve a theoretical2-3 bit resolution improvement to the stepped curve interpolation.

Filter Selection Logic:

Best filter choices are made when program conditions reveal a compromiseproblem. Strong high frequencies, isolated fast transients, andcontinuous low levels encode best with specialized filters. Fortunately,each condition is easy to identify and each is most likely to occur byitself. Conditions such as loud or soft, continuous or broken, andstrong treble are representative of those which are identifiable andcause problems. Program materials are not predictable and solovoice/instruments, synthesizer, percussion, and so forth may presentrapidly changing requirements. Unfortunately, filter lengths, enhancedtransients, filter merges, and identifying program conditions alloperate with time constraints. Therefore, filter changes may have to berestricted when program conditions call for choices which alternate toorapidly.

A best compromise over time is made by memory enhanced variablethreshold smart logic. Normally a filter "call" is initiated in responseto a compromise situation. The call represents the intensity of thedemand for a particular filter, caused by the compromise situation,integrated over a weighted time window. If the program doesn't changemuch and the call is not continuous and doesn't happen again, theprevious best filter remains. Each successive call, its length of timeand intensity increases the response sensitivity to engage that filter.If this choice is made, then the response sensitivity to engage anyother filter is reduced over a preset time and the process like abovenow can repeat for a new filter. This selection method eliminateshaphazard filter toggling and still allows quick filter changes when astrong filter compromise situation occurs.

Although this operation is suited for digital implementation, a hybridanalog and digital circuit is less complex and has allowed easyexperimentation with programs. Diodes, resistors, capacitors,comparators, and current sources make up most of the adaptive decisionmaking elements. Calls are voltages through resistors chargingintegrating capacitors. Call urgency translates to higher voltage for alonger time thereby increasing the capacitor charge rate and amount.Frequency of calls is the number of times this charging takes place. Afilter select threshold is initiated when the capacitor voltage triggersa voltage comparator. To prevent indefinite sensitivity enhancement, atime slot for decision making is created by a negative going currentsource which eventually discharges the capacitor to reset conditions.

Each filter is engaged from a comparator as above. However, whentriggering and a filter change occurs, all the integrating capacitorsfor the other filter choices are discharged and held inactive for apre-determined setup and run time. Once completed the process resets andstarts again without prior memory.

Program conditions are recognized with building block type analogcircuits. High pass filters and peak detectors sort out alias causinghigh level upper frequencies. Peak and average level detectors arecompared and the difference response integrated to identify transients.To reduce the influence of program level variations, voltage controlledamplifiers servoed from RMS detectors are used to scale the peak-averageoperation. Each of these circuit groups then produces a positive goingaveraged output voltage whose amplitude and time duration is related tothe degree of filter compromise, or the desirability of choosing aparticular filter type. Strong high frequencies, fast peaks at variousprogram levels, and high overall average intensity all convert tosimilar filter call voltages, each proportional to the magnitude andrepetitive nature of the program event.

Filter Switching and Merging:

Filters have different lengths, instantaneous phase shifts, time delays,responses etc. Simple brutal switching from one filter to another wouldcreate serious glitches, and other very audible disturbances. Some formof time alignment, fading and merging or a parametric change within thefilter from one type to another is necessary. All of these techniqueshave been done in both analog and digital filter changing. Early lightbulb photo-cell and VCA type fader-switchers are common analog methods.Many digital synthesizers use mix, merge and parametric filter changesin various combinations to produce inaudible transitions. Good examplesare Fairlight and Synclavier machines which have elaborate digitaltracking filters which work on these principles.

A simple implementation would use LED photocells and time delaycorrection to switch filters. For more advanced versions, the filtersare implemented using DSP programs, and DSP programs are used to performthe mix, merge, and coefficient changing functions.

4. Frequency response limitations imposed by industry standard samplingrates:

The frequency response of digital systems is fundamentally limited tohalf of the sampling frequency, in accordance with the Nyquist theorem.For a current digital recording media, Compact Disc, this means that onecannot record anything above 22 kilohertz. This limit was chosen basedon the assumption that the human ear cannot hear sounds above about 20kilohertz. Recent research has shown, however, that humans use transientinformation in sounds with frequencies much higher than that todetermine the direction from which the sound has come, and thateliminating those very high frequency components impairs ones ability tolocate the source of the sound. The inner ear actually has nervereceptors for frequencies up to about 80 kilohertz. Therefore, if the"brick wall" low-pass filter, which is a necessary part of all digitalrecording, removes frequencies above about 20 kilohertz in transients,it reduces the level of realism in the sonic image.

In accordance with the invention, the waveshape of critical transientsis reconstructed at the reproducer based on information sent from theencoder over time. The steady state bandwidth of the digital channel isset by the industry standards, but, for occasional transient events,additional information on the shape of the waveform can be spread out intime and sent along for use by the decoder. There are a number ofdifferent methods which can be used to accomplish this, all of whichmake use of a control signal or "side channel" of information sent alongwith the main signal, described in more detail later. They are allnon-linear processes and therefore should be used sparingly.

The methods of transient reconstruction employed fall into threecategories:

a. Waveform synthesis

b. In-Between sample generation

c. Slew rate compression

All of these methods rely on starting with an accurate waveform of thetransient resulting from an oversampled original signal which has thehigher frequency information intact. In the waveform synthesis method, atransient to be reconstructed is identified at the encoder, and it'swave shape is matched to one of a number of predetermined "standard"transient shapes, which are known to both the encoder and decoder. Acommand code identifying the shape is sent through the control channelto the decoder, which regenerates the shape, either by reading it out ofa lookup table or algorithmically generating it, and scales it to theamplitude of the bandlimited transient arriving in the main signal. Thedecoder then uses the synthesized waveshape to correct the shape of thetransient and approximate the original. The correction can be in form ofa difference between the band limited transient and the original, whichis added to the band limited signal at the decoder. Obviously, only alimited number different corrections can be used, since one must bechosen in a reasonable time at the encoder, all of them must beremembered at both ends, tokens must be assigned to designate thechoices, and time is required to synthesize and scale the correction atthe decoder. Nevertheless, it is possible to achieve an apparentincrease in available bandwidth with only a few shapes. This method hasno compromise to the shape of the bandlimited signal except the presenceof the control command, and therefore is not audible on non-decodedplayback.

The in-between sample generation method is very similar to the above,except that instead of sending a token representing a rememberedcorrection, the encoder sends the actual waveform correction over thecontrol channel, spread out in time to accommodate the low bandwidth ofthe side channel. In it's simplest form, this correction can be thevalue of a single "in between" sample point falling between the normalsamples of the band limited signal. The decoder can use this point tocorrect its interpolation of the signal as it generates an oversampledsignal prior to conversion back to analog form. As above, the onlyeffect on non-decoded playback is the presence of the control channel.

The slew rate compression method is different from the above two in thatthe additional information required to construct the transient is spreadout in time and sent as part of the main signal. The control channel isused simply to activate the process. This method is conceptually similarto a technique used to enhance the apparent bandwidth of a video monitorduring transients by slowing down the scan rate during the transient andspeeding it up again to make up the lost time. When the slew rate, orrate of change, of the waveform exceeds a threshold, it is limited to avalue which can be represented accurately in the band limited signal.The degree to which the speed is slowed down is scaled to the speed ofthe original transient so that the decoder can infer the original slewrate from the slow one which it can observe in the recorded signal andspeed up again. Since the transient is spread out in time, the time mustbe made up somewhere, normally afterwards. In order work properly, thetransient must be isolated so that information near it in time is notlost. This method definitely does have sonic consequences for undecodedplayback, but analog tests indicate that a surprising amount of slewlimiting can be done without being objectionable.

Dynamically controlled system:

All the above improvements are most effective when the reproduce decoderchanges to complement conditions of the recorded program. The recordencode process can generate a hidden control code concealed in noise asone method of controlling these activities. Random modulation offorbidden numbers in an error correction code or user code is anotherway that the control codes can be included with the program data. Thesecan be continuous or initiated when needed. When the code hides in theprogram, digital copies from one format to another will preserve thecode whereas analog copies will not. These features can identifyunauthorized copies, as well as convey production process informationthat might be used for motion picture work, etc.

Algorithms and lookup tables in the decoder provide curvature shapes,time constants, level thresholds, multipliers, filter coefficients andother useful data also "known" by the encoder. Without continuouscontrol information, the system can run default where therecorder/encoder is set up to anticipate the reproducer response. Eitherfeedback from an internal test for best encoding or feedforward of apreviously worked out response will do this. Most control activity isneeded to access and change a particular complementary reproducefunction or correction. Hence the improvement is much greater than theinformation bandwidth loss necessary to make the improvement.

5. Digital to analog and analog to digital crosstalk:

The smallest analog signals, in the 100 microvolt range and under, areeasily contaminated or interfered with by digital data streams havingmillions of times greater energy. Faster processing increases the energyper bit as well as the number of interferences per second. The samesituation occurs with larger numbers of bits. Interconnects, cables, andenclosures pick up this energy, store it, and create delayed compoundinteractions. Higher speeds require smaller packaging which increasessuch crosstalk, unless wires and other parts are also made smaller.

FIG. 5b Analog waveform changes occurring from digital interaction.

FIG. 5c Sample and encoding errors from rapidly changing waveforms.

FIG. 5d Sample and encoding errors from low level waveforms.

In accordance with the invention, a silent conversion system is used toresolve the above problem.

Normally a digital system operates with a continuous clock which timesits internal operations. Millions of timed events occur each second.Hence the system state may change or be in a state of change at anymoment, particularly the critical sample time when high accuracy isneeded most. Sample time jitter and digital to analog crosstalk mayresult.

The system of the present invention stops all operations long enoughbefore sampling to allow the energy stored on cables and other energystorage parts to dissipate. One pulse initiates sampling which thenoccurs during electrical silence. Once signal capture is complete, otherprocesses resume.

A number of prior art approaches have been developed to reduce some ofthe above distortions and are described as follows:

1. Group delay of low pass filter accomplished by using an all passphase shift network.

2. Quantization noise reduction from use of 1 or 2 least significantbits "keep alive" ultrasonic dither.

3. Reduced granularity noise from use of balanced or push pull circuits,achieving high common mode rejection of noise spikes and digital-analogcrosstalk. Further reductions made with optical isolation of logic andconverter systems, as well as high impedance isolated power supplies.

4. Slew induced errors reduced by super fast symmetrical analogcircuits.

These improvements help reduce the harsh, congested sonics, and mayslightly expand spatial sense in good recordings. However, inner detailand correct space perspective are still lost even with such prior artapproaches.

Referring now more particularly to FIG. 8 of the drawings, there isshown a processing system, in accordance with the invention. The systemmodifies signals to achieve 18 bit performance from a standard 16 bitconverter system. As mentioned previously, the average level of verysmall signals is expanded while the occasional instantaneous peaks ofvery large signals are soft limited. When both operations are carefullydone and are digitally encoded and then reproduced in a standard mannerwithout decoding, the sound is improved. Ambience and articulation morelike the original program does occur, even though the process would beaudible without the intervening analog to digital and digital to analogconversion. By the time this expand limit process becomes clearlyaudible on standard equipment, the dynamic range for fully decodedreproduction in accordance with the invention has increased almost 20dB. Average resolution is substantially more than 18 bits.

Each numbered subsystem element in FIG. 8 is an important stand aloneoperation typically performed by an independent circuit card or module.The corresponding schematic circuits for implementing the system of FIG.8, while deemed to be within the design purview of one of ordinary skillin the art, are included for convenience in Appendices A, B and Cincluded with this specification. These circuits are numbered tocorrespond to their respective block functions in FIG. 8.

The small signal average expand subsystem 61 operates in the followingmanner. The incoming audio signal is band restricted to 5 Hz to 500 kHz,thereby preventing DC level shifts, supersonics, and radio frequencycomponents from overslewing amplifiers and influencing process controlparameters. Components C1, L1, R1, C2 and R2 and buffer followercomponents J211 and J271 in Appendix A perform this inside-outside worldisolation. Two active signal paths are provided, one passing through avoltage controlled amplifier (VCA and IC1) and the other originatingfrom the buffer. Both signals are in phase, however, a control signalmarked "compensation input" can set the VCA output from -40 dB to wellabove the buffered output.

Since VCA devices are well-known distortion producers, thisconfiguration allows the clean buffered signal to pass uncontaminatedwhen the VCA is shut-off at it's -40 dB gain. Only during very smallsignal conditions, when distortions are less important, is the VCA gainmade large, allowing its substantial output signal to be added to thebuffered signal. A higher overall output results. Remaining circuitcomponents perform housekeeping functions necessary to prevent crosstalkbetween control signal and VCA output and to adjust for lowestdistortions.

The phase shift network subsystem 62 operates as follows. The bufferedand VCA signals summate or combine in phase at an attenuator R1 R2 R3 inAppendix A. With average program signals in the 0.05 to 0.5 volt (0 to-20 dB range), the VCA gain is set at -40 dB, making its output anddistortions inconsequential to the accurate buffer signal. VCA gainincreases substantially for small signals in the 0.005 and under (-40 to-80 dB range) where monotonicity and discrete step-by-step typequantization errors from the AD-DA process are becoming increasinglylarge. This added VCA output maintains a "keep alive" status or minimumbit number change rate at the A to D converter. Here the signal and itsbackground noise become dither and a minimum useful amount of it ismaintained by the variable gain VCA independent of signal conditions.Sometimes noise dither is added to digital systems and in practice thisnoise is quasi-audible. The active-dynamic dither has similar propertiesexcept that when needed, the original program dynamics can be restoredwith a controlled playback VCA system.

Four sections of all pass phase shift correction circuits and a secondbuffer are placed in the signal path. These stages marked A, B, aretwice iterated. Each section of J557, J211 and J271 is connected to makea unity gain buffer--inverter giving in phase signals at R5 and 180degrees out of phase signal at R4. Both signals combine through Ra andCa making a flat response all pass system having near 0 degrees at lowfrequencies and 180 degree phase at high frequencies. The four combinedsections still has flat frequency response yet exhibit an abrupt 720degree phase shift in the 5 kHz to 30 kHz region. This corresponds to a400 uSec group delay change which partially cancels a sudden group delayshift occurring with elliptical low pass filters. Without compensation,serious transient ringing would ripple modulate on and off thesubsequent peak level limiter and would cause excessive gain modulationat the reproducer restoring or expanding the signal peaks. The groupdelay filter has very little ringing and allows much more predictablepeak limit and expand operations.

The low pass filter subsystem 63 (701 filter) operates in the followingmanner. This is an essential and very troublesome part of all A to Dconverter systems. It stops or rejects frequencies near and above theNyquist limit or 1/2 sampling rate. For idle systems, its "stopping:action must be better than the least bit resolution, while its passaction must be ripple free and in proper phase alignment in the 15 Hz to20 kHz range. At 44.1 kHz sampling and 16 bit encoding for Compact Diskformats, the filter must drop at least 85 dB between the 20 kHz audiolimit and the 22 kHz Nyquist limit. Mathematics of traditional filterdesigns require compromise decisions related to numbers of parts andtheir signal degradation, and alias, group delay, and ripplecompromises. For the system of FIG. 8, a compromise of more parts forbetter transient and group delay is taken to allow better peaklimit--expand operation. The delayed high frequency part of a sweep,which happens first, can add or subtract from lower frequency parts ofthe sweep happening later. Thus, the instantaneous frequency responsecan change with fast changing signals. Ripply beats throughout envelopeand "tail" occur when such a sweep repeats back to highest frequency. Ashortened envelope also occurs as delayed high frequencies ofuncompensated filter occur within the sweep envelope. In this regard thedelayed high frequencies continue ringing and internally reflect withinthe filter after the sweep envelope has completed and started the nextcycle beginning well above the filter cut-off.

The high level peak limiter subsystem 64 (limiter, expander, dithergenerator) operates as follows. IC101 in Appendix A receives the filteroutput and completes transient response ripple compensation previouslymentioned. Q1 and Q2 with IC 102B perform the peak limit function whileQ3 and Q4 with IC 103 perform the restorative peak expand function atthe reproducer. Added parts IC102A allow one to observe the signalwaveform peak that has been limited and IC104 generated supersonic andnear supersonic noise to dither the A to D conversion.

Completion of low pass filter compensation includes a combinednotch--peak circuit around OP amp IC103. A tuning of 18.5 kHz high Qpartial notch and a 21.5 kHz sharp peak become added filter sectionswhich help smooth and reduce ringing. A rougher, but acceptable,frequency response is made and the peak transient ripple is less than 5%with completion the same as the square wave risetime.

Divider R1 and R2 sets gain structure and source impedance to idealizeddiodes made from transistors Q1 and Q2. These can be "supermatched"pairs having many devices random connected on IC substrate yielding nearideal junction performance. Circuit elements include simple transistors.These behave close to the ideal logarithmic junction relationship of:##EQU3## giving change of forward voltage to operating current ratio.Boltzman constant; temperature, and electron charge are consideredconstant. Instantaneous resistance of dV/dI (rate of change of voltageto current) becomes related to 1/current once limiting action begins.This relationship tracks over a 40 dB (100 times) range for reasonablygood transistors thereby allowing easy record--play peak signal trackingin the 10 dB one to two bit process range. A practical setting is a 2.4volt peak to peak triangle wave compressed to 1.2 v peak through Q1, Q2yielding a 3.5 v pp output at IC102B. This 1 bit (6 dB) compression canbe monitored at "test" output showing the clipped portion which forset-up adjustment is made symmetrical with the 50 k ohm control.Restorative operation is demonstrated by connecting IC102 out to IC106in. Observations at IC103 "protect" output shows tracking ranges inexcess of 20 dB when needed.

Operation of the analog to digital conversion subsystem 65 and digitalto analog subsystem 66 is as follows: Output reconstruction from the Dto A signal must occur prior to the low pass filter subsystem 68,otherwise phase shifts would alter the signal waveshape and consequentexpand threshold points. Consequent sampling feedthrough currents, andinterferences as well as step sampled date require fast circuits.Amplifiers have additional stabilization and speed enhancement.Components marked Rs Cs perform these operations and are specific to theamplifier types used. With decoded step waveforms, the limiter functionmust quickly settle to each level and associated amplifiers must notovershoot, ring, or have unsymmetrical rise and fall times while doingthis. As noted, limit--expand functions must occur at direct coupled orDC pass circuits not having phase shift. A to D inputs and D to Aoutputs satisfy these requirements when the low pass filters are notincluded in the path. In practice, limiting will create upper harmonicsin the Nyquist range which could create alias noise which would confusereproduce reconstruction and add considerable distortion. Fortunately,practical operation allows modest compression and expansion ofoccasional peaks which happen in music and speech program material.Unlike instrumentation signals of constant amplitude maximum energy toband edge character, upper music frequencies are usually harmonics ofless energy than fundamental tones. Alias foldover is then infrequentand occurs only at peaks which best mask these problems. FIGS. 6a-6fshow various signal waveforms during the limiting and reconstruction ofan illustrative triangle wave.

The low pass filter subsystem 68 operates in the following manner. Thepeak reconstructed sampled D to A signal output of IC103 is routed to asimple low pass filter. The 44.1 kHz and up step components are removedand the waveform is rounded and smoothed inherent to the filtercharacteristic. Noise and transient spikes are reduced to tolerablelevels to prevent overslewing the VCA portions of the low level signalcompress circuits to follow.

Operation of the small signal average compress buffer and VCA subsystem69 and Line Amplifier Subsystem 70 (small signal compress, line drive)is as follows: Both subsystems serve similar functions to theirsubsystems 61 and 62 counterparts in the record section. Buffered andVCA output voltages are similar. However, this time the VCA output issubtracted. As before, lower distortion is achieved by operating the VCAat -30 to -40 dB level relative to normal level signals. The output lineamplifier is connected at differential input single ended output.Increased VCA gains reduce signal outputs until at +10 dB, a null or 0signal maximum compression occurs. With this arrangement, any reasonablesignal expansion can be compensated and the system distortion is lowestfor the most probable average level signal conditions.

Control signal generation is accomplished by the limiter-buffersubsystem 71, bandpass filter 72a and RMS detector subsystem 72b andgain subsystem 73. High level peak limit--expand thresholds and lowlevel average gain set controls are needed. Circuit settings allow bothtypes of control to be tested independently of each other. Bestoperation occurs when the control signal anticipates the programwaveform to be processed and, hence, an audio pre-delay is used to allowcontrol stabilization prior to circuit action. Low control bandwidth isneeded to minimize non-signal least bit activity. One method of doingthis is to have active--inactive control status. Since high level--lowlevel program signals do not occur simultaneously, the reproducer canmake its own decisions as to where the control is applied. The unusedoperation then returns to its inactive or nominal process state. Highlevel, normal program type signals, where expand--compress functions areunnecessary, then have inactive control status. As signal levelsdecrease, an internal limit diode/clamp releases and the VCA gainrapidly increases to create summation signals.

Further program level reductions modulate the VCA gain in acontrollable, predictable manner needed to maintain digital "keep alive"LSB (least significant bit) activity. For most conditions, these lowestlevel signals will be mid-band acoustic noises and numerous types ofelectrical noises. The latter may include RF interference, light dimmerpulses, security system signals, and high frequency peaked electronicnoises. These often have low audibility compared to midband acousticsounds. Hence, a sharp cut-off bandpass filter and very wide peak toaverage level capability RMS detector are used to assure that thecontrol signal tracks audio sounds and not inaudible interference.

In the small signal average expand subsystem 61, to handle the entirelarge dynamic range of modern program materials would require very lownoise filters and detectors of quite difficult electronic design. Sincethe average level circuits are low program level active only, theprocess gain can be very high. This allows reasonable circuit voltagesto occur during quiet passages. Strong signals normally creatingoverload are smooth limited with a semi-logarithmic curve to createminimal compression harmonics. Noise, transients, and other uncontrolledoverload behavior are then prevented from crosstalking to the signalpath. Component IC1 is configured as approximately 100 times smallsignal unity gain. Successive diode conduction with increasing voltageuntil the stage has less than unity under signals and maximum outputrarely exceeds 6 v peak to peak at maximum program levels.

The bandpass filter subsystem 72a prevents low level inaudibleelectronic noises from modulating the average low level process. Thebreadboard part covers a 200 Hz to 5 kHz range and is made from twosections of active combined low pass high pass feedback type filters.These have a slight rise at band edge frequencies followed by a near 24dB per octave cut-off. Note, that a front end buffer (J211 and J271)prevent filter impedance loading from interacting with input signals.

The filter output drives an RMS detector module, a DBX-type componenthaving an averaged DC logarithmic output relative to AC input. Asconfigured, a 100 mV output change occurs for each 20 dB input signalchange. This gain structure from the limiter and filter through theconverter gives 100 mV control range with very little noise for themillivolt type signals occurring at least bit resolution limits. Levelson either side of this represent front end electronics noise and normalsignal operation.

Normal compressor--expander compromises are employed to assure minimalVCA gain modulation (distortion) from AC components in the controlsignal. Components C1, R1 perform this response averaging for theturn-off time constant. A much shorter turn-on time constant from theRMS module internal impedance and C1 occurs to allow fastest response tosudden program level increases. The short--long time constant action istypical of compand systems and because of low frequency distortionrequirements, is set very long. (20 mSec on, 500 mSec. off for 20 dBgain change) This very slow response necessitates input signal delay toallow control signal buildup before sudden signals occur when VCA gainis maximum at low level signals. In practice, the delay addsconsiderable distortion, and would not be used in its analog form inhigh quality systems.

Digital processors can perform all of the above level limits, bandrestriction and detection. The needed time delays to get bestperformance are simple, first in first out type, operations. Twoadvantages of long constant operations occur. Low frequency distortionis reduced and control signal bandwidth is much less, thereby reducingthe amount of bit borrowing needed to pass the control through the audioencoding.

The DC offset and gain adjust subsystem 73 is the control signalamplitude, offset, and limit nerve center. It adjusts for tolerancesbetween amplifiers, delay lines, VCA's and the RMS detector and, hence,performs general circuit housekeeping functions. It also is a limiter togive maximum and minimum VCA gains needed to implement the controlsignal inactive and minimum input signal presets.

For system considerations, both VCA's and the RMS detector operate withlogarithmic control to signal and signal to control relationships.Hence, changing an offset changes a fixed program gain in dB. This makespossible large dynamic range gain control and yet still maintains lowlevel signal control. Both can occur with reasonable control signallimits. In addition, control system gain changes, such as limiting, givedirect dB to voltage gain ratios making very simple compress--expandratios and assuring input--output tracking by simple polarity reversalof the control signal.

The offset gain adjust circuit inverts the control signal for inputoutput tracking, adjusts DC offset of each to match gains at apre-determined control signal level, and has control gain adjusts tomake program level increases match reproducer gain decrease for varyingcontrol levels. Diodes CR1 and CR2 perform level limits to preset amaximum practical expand ratio and a maximum compress ratio underplayback. As constructed and configured, this circuit hascompress--expand ratio adjustment interactive on a single control andthe threshold of when the process starts on a second control. Sincethese are DC level of dB operators, process control signals to thesepoints will create dynamic process changes. At present, this is a manualadjustment. However, the process start level can be change dependent onprogram activity and other considerations to reduce audibility withnormal non-process playback.

The delay line subsystem 74a&b is a balanced DC coupled self-clockingvariable delay line. Charge coupled devices are operated push pull withstaggered clocking to make the lowest possible DC drift, distortion, andclock noise from relatively poor performance devices. DC to 25 kHzminimum overshoot, 80 dB dynamics are achieved.

One delay 74a is used to allow control signal stabilization to preventVCA overload from sudden signal changes. This delay also allowsanticipatory process control signal strategy to be computed. A secondline 74b is for test purposes when using a system without the hiddencode subsystem to match the A to D and D to A encode-decode process timeto make output signals track inputs. Bit borrowing, in which the controlsignal is noise encrypted and hidden in the least significant bit orbits of the digital signal, is the normal mode of operation of thesystem.

Limiter dynamic control via VCA input/output tracking can be had at highsignal levels by removing the limiter-buffer subsystem 71 and operatingthe system as a straight compressor-expander. Since the peaklimiter-expander is within this system, its operation is changed alongwith the gain variation programmed with the offset gain adjust subsystem73. As noted before, a process control input can be operated so that alltimes when signals are loud, a certain percentage of limiting takesplace. This is program dependent as some classical music will haveinfrequent peaks while studio processed rock and roll is more likelyhard limited and will have many small peaks occurring frequently.Although this threshold control may be accomplished manually, numerouscomputed variations will work more effectively to keep the process leastaudible when reproduced on a standard non-restoring player, just as inthe case of the low level process dynamic control. As with the averagelevel expand-compand, the limit expand threshold control need only bevery low bandwidth. A 10 Hz control bandwidth is adequate and since onlyone operation occurs at a time, only one control for both operations isneeded. The player can determine program level and switch functions. Inthe illustrative system described, the control is manual since eachoperation is a different set-up. However, there is no foreseeabledifficulty in making this automatic if duplicate VCA's were set up forlimiter gain structuring.

The following further describes the theory, design concepts, and earlydevelopment, construction and operation of an encode-decode systemdemonstrating the basic principles of the present invention. Itsoperation is analogous to the reconstruct process based on choosingoptimal curve fitting techniques to get best waveform reconstruction. Asdescribed, the process changes for different signal conditions and thenumber of such process optimizations per unit time can range from a fewper program to many times per second. Even faster operation changes arepossible. However, the control signal needed to access the properoperational program becomes more complex and bandwidth consuming.

The basic system contains two record processors and two decodeprocessors, each of which is complementary to and matching as a system.Either system is selected automatically by variable resistancephotoconductive cells working as slow faders in the signal path. Lightemitting diodes illuminate these cells and are driven by variable levelcontrol signals emanating from a signal analysis logic circuit. Duringoperation, the logic chooses the least distortion process based onsignal conditions. Similar switching and routing can be accomplishedwith voltage controlled amplifiers, digital attenuators, and fieldeffect devices or other components acting on the analog signal. Similarrouting, mixing, or merging operations can be made from digitalprocessors operating on numbers representing the signal. Such operationslike those with the photoconductive cells can be transient disturbancefree by virtue of their slow switching action. Each record process, andits complementary reproduce process resembles a filter-equalizeroperation which is made optimum for the program signal. Both partsoperate as a system so that the encoder can anticipate reproduce errorsand can create complementary corrections. In this manner, the record andreproduce circuits are not working as an individual, stand-alonetheoretical ideal, rather as an optimal system. The breadboard designhas two such systems, one for best articulation and transient responseand the other for lowest distortion.

Filters, equalizers and curve fitting operations are accomplished asfollows. One can define a filter mathematically by coefficient stringsin polynomial sequences. In addition, the same filter can be defined byhow it responds to a given waveform stimulus. Essentially, curve fittingin time, frequency domain, and amplitude is created from numbers whichcan be stored in tables. For analog systems, such operations areperformed by circuit elements connected to band restrict, equalize, timecorrect, and to perform dispersion operations on a signal. Thesecircuits can also be analyzed back to similar polynomial coefficientswhich can run as digital process programs. As can be seen, very awkwardcircuit construction problems occur when one must alter these numbersfrom time to time, as would happen with a dynamically changing process.Multiple component values of inductors, resistors and capacitors, aswell as gain stages, would be all simultaneously changing to producesuch a merge operation. This can, however, be accomplished by digitalprograms. Dynamically changing digital filters have become practicalonly recently as the necessary processing power has become economicallyavailable. Of course, simple networks such voltage controlled parametricequalizers and variable RC section tone controls have been available fora long time. However, complex variable filters are still rare. As onecan see, the dynamically changing curve fitting operations can behandled most directly by digital processing. In an analog systemcomplete filters must be changed, whereas the digital process merelychanges filtering. Both have similar potential curve fitting capability,however, they differ during the transition region from one process toanother.

In theory, only one set of filter coefficients is needed to make a nearideal analog to digital conversion and its reciprocal, provided allsignal activity in the frequency domain is greatly removed from theNyquist sample limit and there are ample numbers to characterize thesignal. Commercial digital standards do not allow either of theseconditions and, consequently, some practical state-of-the-art compromiseof time, transient, alias, quantization, and flatness of response mustbe made. The best of each performance aspect cannot occur simultaneouslyand one of ordinary skill must choose a compromise based on knowledgeand subjective experience with audio programs.

As previously indicated, the signal to noise ratio of digital processescan be excellent while complex distortions at average signal level canbe higher than with good analog systems. For high quality work, there isa need for improved resolution, time and transient accuracy as well asreduced high order distortion. This aspect of the encode-decode system,in accordance with the present invention, addresses such a need.

At this point, a further understanding of digital distortions will proveuseful. Typical digital systems have between 0.01 and 0.05% totalharmonic distortion (THD) at high signal levels and about 10% cumulativeerrors in the time-transient domain. Most analog systems have oppositeproblems to these, as they often operate at above 1% THD but seldom havemore than 0.1% transient time error. Under low to average signalconditions, digital THD increases while analog THD decreases. As noted,digital distortions tend to be upper order and non-harmonic and,therefore, stand out due to their non-musical nature. Analog distortionsoccur less frequently and are less objectionable, even at higher levels,as they tend to blend in or musically merge with the signal. Similarproblems occur with transient time domain type distortions. At first, itwas thought such problems were inaudible, since simple square wave testswould show few sonic consequences from such distortions. Today, we canshow serious deterioration of spatial sense, as well as lost innerdetail perception as a result. As digital time domain distortion is muchmore complex than the simple ringing measured in early tests, resolutionperformance of industry standard 16 bit encoding is also inadequate. Asystem which can produce 10 volt peak to peak signal will haveapproximately 150 uV best possible resolution from one least significantnumber step to the next. Practical systems have signal discontinuity of4 to 8 times greater than this, as the state-of-the-art has not yetallowed near theoretical performance. A 20 to 50 uVolt discontinuitylimit is typically considered just audible. Practical systems havedistortions often ten times higher than this.

As noted earlier, the peak signal limit-expand and low level averageexpand-compress operations deal with resolution problems. Otherdistortions from time shift, alias, and quantization, which are inherentwith even ideal encode decode operating to industry standards, stillremain.

Distortion reduction may be accomplished in the following manner. Mostdigital distortions can be predicted, as they are strongly related tosignal conditions which are easy to identify. It follows that, for agiven signal, one can choose a best encode-decode process having theleast audible or sonically damaging distortion. If one must operate toindustry standards where the Nyquist limit is just outside the audiorange, then a transient response versus alias compromise exists. Thiscompromise occurs when requiring flat passband response and a verynarrow pass to reject transition bandwidth. As the signal changes, onecan choose the best process.

In practice, phase and time response are not equal from one complexfilter-equalizer network to another and a slow fade or merge is neededto prevent inevitable switching transients with process shifting.Similar problems are dealt with for analog noise reduction processors.With digital processing, these operations of merging from one optimumfilter or curve fit to another can be lookup table coefficients whichare accessed as a sequence to merge from one filter type to another.While phase anomalies still occur, the decoded signal can, however, befree of beats or cancellations which plaque analog fader type systems.The mix or filter change merge occurs just fast enough to preventaudible transients or other parametrically generated phase disturbance.

Since digital process timing is almost always crystal controlled, therecord--play transitions can be made to track each other by pre-timedsequence programming which can be initiated by a single command. Thiseliminates the need for continuous numeric control and higher bandwidthfor the control signal. All process types, transition speeds, andintermediate coefficients can be stored and run as a program from asingle, one time command and the recorder-reproducer are effectivelylocked to each other.

The basic analog system uses resistor-capacitor time constants withinthreshold sensing logic to simulate pre-determined transition speeds andthe resultant tracking of intermediate filter mix states. In addition,other time constants also serve as internal memory to add hysteresis orhold back to decision making operations. These allow a first time quickprocess change decision and a reduced sensitivity to further changesthereafter until the time constant resets. This prevents unnecessaryprocess changes during grey area or uncertain signal conditions. Like adigital system, the analog system has the ability to operate with simpleswitch on switch off control where the output tracks the input and wherethe output or reproduce subsystem does not have to detect signalconditions to do so.

Normal analog systems are not DC or direct coupled as is the case withdigital, and these would require an additional data channel with alinear control signal or an internal analog signal conditions detectorto make such a system operational. For practical operation, the basicanalog system has been tested without buried or hidden code control anda third control channel with appropriate time delays has been included.

There are numerous ways to sneak through and hide the control signalwithin digital systems. As previously pointed out, one can random noiseencode-decode a control signal in the least significant bit(s). Thisoperation has no counterpart in the analog domain as it is nearlyimpossible to locate minuscule portion of a complex waveform carryingthis information. In the digital realm, the least significant bitactivity is always known. Hence, this bit can actually be borrowed forcontrol purposes. Other ways to hide a control signal include usingforbidden numbers or unused data words or number strings within adigital system which the same or a different system considers an erroror nonexistent data. When the forbidden numbers are carefully chosenduring encoding, the reproducer will recognize the error, but stilldecode its data signal correctly. Of course, the forbidden number isdata which can be extracted and used for the control function. Eithermethod of concealment has enough bandwidth within Compact Disc standardsto allow ample bandwidth or information carrying capacity to passcomplex control signals. Any signal degradation from performing thisoperation is very small when compared to the improvements resulting fromthe added process power and re-construction capability being controlled.

Present industry standards are largely based on providing goodperformance in terms of flat response, low harmonic distortion, and highsignal to noise ratio. Time, transient, alias, and resolution arecompromised, but their problems or deficiency occur predictably withsignal conditions. Consequently, a control logic must analyze theincoming program material and determine a best process.

The relatively simple basic analog system circuit makes quite accuratedecisions of quantization versus alias distortion baled on highfrequency intensity and its ratio to average program level. This followsfrom the flat frequency response-sharp cut off compromise of the lowpass filter design. In practice, the filter must be 85 to 90 dB down atthe end of its 2 kHz transition region. Just prior to this, it must beflat to 20 kHz. Unfortunate and serious transient ringing must occur, ascan be demonstrated from the analysis of a square wave with its upperharmonics sharply removed. The filter having good transient responsewill not remove enough alias causing upper frequencies.

Live program spectral energy, in the transition region and above, isunpredictable and ranges from bursts caused by microphone element peaks,instrument overtones, amplifier distortions, etc. Hence, a simple highfrequency level detector can determine whether added filtering for aliasreduction is needed or not. Since these distortions can be covered up byprogram sonics, an added weighting factor of reduced rejection duringhigh program levels can be used. Therefore, the detector looks forratios of high frequency Nyquist energy to average program levels todetermine when more aggressive filtering is needed.

Essentially, a reduced second derivative around the cut-off slope yieldsimproved time and transient performance. It is assumed that symmetricalfilters having constant group delay are used, as they are practicalanalog and digital process types. These have symmetrical pulse behaviorand can be made to mix/merge from one curve fit shape to another withoutaltering group delay and creating excess phase interference duringtransit time. Practical systems can have as much as 200 uSec. timeshifts near cut-off when full 90 dB alias rejection occurs. Thesenumbers relate to about 0.15 inch rapid displacement or doppler shift ofhigh frequencies which can occur very rapidly with music type waveforms.Certain types of transient intermodulation distortion (TIM) can occurunder similar conditions. When corrected to 3 uSec/kHz change of upperpassband conditions, a filter may have less than 50 dB rejection.However, as can be seen, a best choice compromise switchable system ispractical.

A second group of compromises relates to quantization distortions andthe smallest signals which can be processed. As noted, level changeoperations reduce these problems in a compatible manner. Some furtherimprovements can be made by anticipatory forced resolution enhancement.Like the alias/transient operations, these are also curve fitting innature and can be accomplished by record-play circuit systems resemblingequalizing filters. In this case, a forced high frequency extensionduring record is made when signals have small amounts of high frequencyinformation. When normal signal levels with high frequency content arepresent, the frequency response of the system is flat, but when thesignal level is low and there is little energy in the high part of thespectrum, the frequency response in the record half of the system isboosted. The play circuit does the inverse operation. The overall leastsignificant bit activity is substantially increased and more informationbecomes encoded via duty cycle modulation and increased dithering. Whenthe record equalization (EQ) contour rises very sharply, most of thisadded information is just at and slightly beyond audio range. It haslittle effect on standard players or on hearing, because hearing acuityis low for these small, low level signals. Essentially, one has tradednon-harmonic distortion for a similar amount of harmonically relatedprogram distortion. To a degree, the less accurate the player, thebetter this process works to disguise grainy noise as upper musicharmonics.

Of course, a decoder can be instructed to perform flat responsereconstruction and there would then be more data bits making thecomplete signal. Hence, quantization noise is reduced. This is anothercurve fitting operation which might be called dynamic dither, as it mustbe removed in the presence of strong signals. If left continuously on,alias or beat frequencies will occur from strong signal harmonicsinteracting with such excess energy high frequency dither. Clearly, theprocess must shut-off under intense high frequency conditions whereresolution benefits become minor.

Control signals for resolution enhancement and distortion reductionprocesses can be derived by looking for critical energy in the aliasfrequency range. High ratios of these high frequencies to average signalconditions are indicative of possible foldover distortions made audibleas they are not masked by program material.

Most significant are complex high frequencies by themselves such asthose encountered with cymbals since low frequency problems arecompletely unmasked and are audible 60 to 80 dB below where hearingacuity is strong. Such signals and how fast they change can be sensed todetermine a best process. Quick high frequency bursts above averagelevel conditions suggest least filtering and best transient response,provided some midband energy in the predicted alias range is present.Low levels of high frequency energy suggest quantization or dynamicdither correction.

Since some process/filter/equalizer coefficient change operations can bemore audible than others, some maximum number of changes become part ofthe decision making process. Dynamic dithering and resolutionenhancement (EQ) are simple high frequency operations which can beturned off and on rapidly without sonic consequences from sudden phaseshifts, beats, etc. Transient alias switching is much slower astime-phase changes do occur. Because of these possible process toprocess time change constraints, it is necessary to look ahead toobserve the before and after signal conditions surrounding the decisionpoint. In addition, the occurrence frequency of these changes, past andpresent, is important to prevent process hunting or decision instabilityresulting in unnecessary process changes.

Circuits to perform decision making are deceptively simple compared towhat one might expect from the aforedescribed functional descriptionsand the same holds true when the circuit equivalents operate fromdigital systems programming. The basic analog system uses analog circuitsubsystems to perform these operations. These include, delays, voltagecomparators, spectral analyzers, multipliers and signal detectors withtime constant memory. High frequencies are detected and routed to threevoltage comparators. One is set to detect minimum HF thereby switchingon dynamic dither. The second is set to maximum allowable HF to switchon the large anti-alias filter. The third has a variable thresholddependent on program level. Each comparator has its own time constant orhold-off, so that, once fired, or its on-off state is changed, then acertain time must lapse before the circuit will respond again. Inpractice, these time constants are performed by diodes charging resistorcapacitor networks and, as configured, the charge to discharge time canbe easily made unsymmetrical. This behavior allows quick decisions of a"one shot" nature without having the circuit jump from state to statefrom near threshold conditions. Lamp/LED sources illuminate signalsteering photocells to give quick fades from one process to another. Aswith the comparators, the lamp drivers for each process type havedifferent on-off time constants to accommodate time-phase differencesfrom one process to another. In addition, several time delays are usedto allow logic action to happen prior to the signal conditions requiringthe change.

Unlike analog noise processors which require record-play tracking andvery carefully worked out signal thresholds, the basic analog systemprocess decisions can be very rough. Accuracy is unnecessary as thereproducer process is always tracking. Since the operations are industrystandard compatible, no major disaster will occur from a wrong decision.Hence, the analog circuits in the basic analog system have worked "asis" without refinement.

It is clear that when a digital signal and process is used, the encodingmust have greater accuracy and resolution than the final industrystandard product. One method to assure this is to encode with added bitsat a high sampling rate and then perform successive decimation andarithmetic roundoff or truncation to make the final format. (44.1 kHz,16 bit) Processing becomes multiple stages of delays, filtering,equalizing, instantaneous gain changing and averaged gain changing. Thesignal is analyzed and the results are used to interrogate a process"rule book". Several processes and their reproduce conjugates areavailable to be chosen based on predicted error and best signalreproduction. Once determined and initiated, transition parameters areaccessed and the process starts changing. During this decision makingtime, the music signal has been delayed to allow the process transitionto complete prior to the signal being matched. A control word isgenerated and encoded for inclusion in the recording, so that thereproducer can access from its memory the complementary process and itssynchronous transition parameters. Both operations commence relative totheir timed sequences and to their stored data. Since the recorder hasalready simulated the pre-programmed reproducer action or thecorrections for consequences of its action, both processes changesynchronously within the time accuracy limits to the encoder-decodercrystals or clocks. The system then changes itself without having majortransition aberrations and then operates with the best performance forthe signal conditions.

Referring now more particularly to FIGS. 9 and 10 of the drawings, thereis shown a system which can choose an optimum recording process and itsreproduce conjugate to achieve low alias or quickest changing, fastestsettling transient response. Phase interferences during transition timeare controlled by "fader" time constants and signal delays. The logiccircuit has one comparator set up to change state when alias distortionwould be greater than an approximate 40 dB below peak program level.

This circuit, shown in Appendix C included with this specification,contains four sets of back-to-back LED-Photoresistive cell switchers anddriver circuits. Signals are delayed to allow process decision times andtransition time constants are adjustable to allow smooth fading betweentwo process signal paths. Controls are derived from an analysis filterand detector made sensitive to highest alias frequencies. A peak leveldetector sets a voltage threshold from which a comparator can bereferenced. Alias components above this level setting will switchprocesses. A second program delay is used to synchronize record-playtracking and effectively matches variable time constants of the LEDdrivers. Different control settings allow this circuit to operate eitheralias/transient as a dynamic operation or quantization/distortion as anindependent operation.

These circuits are set up to be compatible to industry Compact Discstandards. The encoded product having these variable-dynamic processeswill play back with equal to or better sonics than without processing,even on a standard home player without the decoding features of theinvention. Circuit subsystem blocks correspond to those used inpreviously discussed embodiments of the invention.

As observed in FIG. 11, "process A", there are waveforms and distortionplots starting with input signal, output of uncompensated encoding, theconjugate restorative response, and the overall corrected systemresponse. Test signals include slow sweep forward from 20 Hz to 30 kHz,3 kHz square wave, and distortion measured from 20 Hz to 30 kHzfrequency sweep at near operating level.

FIG. 12 illustrates "process B" and uses the same format as FIG. 12 for"process A", except the plots are for the fast transient process.

One method of overcoming the frequency response limitation imposed ondigital recording systems by industry standards and its effect ontransient response is the use of slew rate compression, as discussedearlier. Slew rate limiting and expansion operate in a similar manner tothe peak amplitude methods previously described. As before, a nonlinearelement is introduced in the signal path to perform the desiredlimiting, and the expansion or reconstruction method involves placingthe same device or circuit in the feedback loop of an operationalamplifier. Variable conduction of diodes with increased voltage is usedfor peak amplitude limiting whereas increased current through acapacitor with increased signal speed is used for slew rate limiting andexpansion. Slew rate limiting takes an event and spreads it in time and,hence, its use must be limited to occasional events like those occurringin musical programs.

The basic system using an analog implementation of slew rate compressionis shown in FIG. 13. Typical waveforms associated with its operation areillustrated in FIG. 14. Schematic diagrams of the key modules areincluded in Appendix D included with this specification and arediscussed below.

Fig. D1 shows an example real time slew limiter having a circuitconfiguration somewhat analogous to the diode limiter types previouslydescribed. Here a representative wideband square wave signal withtransitions of many volts per microsecond is shown applied to anamplifier, marked A₁, which is constructed to have a restricted slewrate performance of much less than one microvolt per microsecond. Itssquare wave output now has a well defined rise and fall character. Whenthe input and output of this amplifier are compared and the gainstructure is appropriately set to cancel slowly changing signals, theslew rate limited part of the signal becomes available. A bridge-likecircuit of the amplifier A₂ and the resistors R₁ through R₆ perform thistask and its output is the distorted portion of the signal occurringduring slew limiting. When this correction signal is appropriatelyamplified and added to the slew limited signal, the original inputsquare wave is restored.

A very high performance slew limiting amplifier is needed for this taskand a specialized circuit configuration must be carefully worked out toprevent sub-harmonic, recovery, and overload distortions. In addition,the degree of slew rate limiting with respect to signal speed must bepredictable so that acceptable reproduction can be reconstructed whenthe correction signal is not present, as might be the case in a simplereproducer. A standard operational amplifier will not work adequatelyfor this task. Fig. D2 shows a simplified conceptual variable slew rateamplifier where all parts operate in linear class A so that conductionoccurs under all signal and limiting conditions. Voltage controlledvariable current sources marked I+ and I- are used to achieve slewlimiting. Two of these circuit groups marked A and B oppose each otherand the balance between them is modulated by the input signal throughFET devices marked C and D. Current limits either side of balance arerestricted by diodes E and F which are in turn controlled from abalanced phase inverter FET marked G. Slow changing signals create smallcurrents through the capacitor marked H and have inconsequential effect.Large fast changing signals demand more current and the limiterrestrictions then restrict slew rate in an ever increasing mannerfollowing the diode conduction versus voltage curves. Hence a lowdistortion predictable and controllable symmetrical slew limit occurs. Amore detailed schematic is shown in Fig. D3.

In practice, it would be desirable to be able to reproduce an occasionalfast signal such as percussive transients. These may have large fasttransition waveshapes which are quicker than filter and samplinglimitations allow and the above circuits arranged like the diodeexpander will perform this operation without requiring an externalcorrection signal. Fig. D4 shows this arrangement. The variable slewamplifier is now made slower than the anticipated input signal from thereproducer so that the difference between the recorded signallimitations and the reproducer amplifier performance now becomes asynthesized correction signal. As before, R₁ -R₆ and amplifiers A₁ andA₂ are like a bridge which cancels unlimited signals and presents theslew rate difference between input and output. Previously, the limitedand corrected signals were added to restore the input. Now anovercorrection is made to anticipate the signal that would have been atthe encoder input prior to band limit filtering. This operation thenuses an overcorrection signal which will vary from one signal conditionto another, hence a controlled variable gain device, VCA, replaces thefixed R₇ of the previous circuit. When the control signal has beenproperly set up for this event, an error correction signal can be addedto the input signal to yield a much faster transient reproduction whichnow more closely resembles waveshapes of wider bandwidth input signals.As can be seen, a transition shift approximating slew restoration occursand if time integrity is needed, the signal must be advanced back intime by a variable delay so that during this reconstruction, the edgetransition occurs where it would have in the original program material.

Both slew rate and correction signal gains are controlled. These areanalogous to curve segment shapes which might be stored in and recalledfrom lookup tables and size scalings which can be determined fromexamination of the signal. Capacitors and diodes from analog circuitscreate predictable slew dependent curve shapes and voltage controlledamplifiers respond to size information. Rate of change of numberscompared to some averaged number scale and multiplier coefficients inmemory simulating curvature or a non-linear function can do the sameoperations in the digital domain. Either operation depends on havingfirst tested the record and predicted reproduce synthesis duringencoding and then generating a control signal which sets up thereproducer to track the best tested results. To do this with the analogcircuit, the input signal is lowpass filtered, possibly slew limited,and then compared with a yet lower slew rate limit circuit to get acorrection signal. VCA gain is then adjusted to get a best match betweenband restricted and unrestricted signals. Slew limit, VCA settings, andengage time become the control information. Since the reproducer has thesame circuits as those used for encoding, the output waveform willtrack. Clearly many other analog and digital methods to determine andsynthesize slew limit and expand can be used. However, a unique aspectis that these operations can synthesize many high definition points of awaveshape portion from pre-coded curve shapes which are accessed from alow information content control signal or can perform a first orderapproximation of the waveshape in the absence of control.

A presently preferred embodiment, in accordance with the invention, willoperate primarily in the digital domain and has the same very basicoverall system design, as shown in FIGS. 3 and 4 of the drawings. As inthe case of the analog system (e.g. FIG. 8), each subsystem is asubtantially self-contained circuit or functional module which performsa unique operation. Input and output signals of these modules are quiteoften similar from one design or product to another. Hence, if thecomponent or subsystem added to improve performance does notsignificantly change these intermediate signals or the circuitconfiguration, then compatibility to standard equipment and recordingsis much more likely. In our case the "DSP" or digital signal processingsubsystems in 100-102 and 104/105 are the unique elements while theremaining components of the system have few changes and are left as theynormally appear in products.

As best observed in FIG. 15, a more detailed diagram of the presentlypreferred digital embodiment of the encode system, highly specializedoperations are performed by functional groups of electronic components.Quite often, each performs a unique task which can be examined,evaluated, and described independently without involving other portionsof the digital system. Hence, each element is a functional buildingblock much like a sound system component which can be specified andcompared to others.

The analog input signal is applied to a balanced input amplifier 201followed by a supersonic low-pass analog filter subsystem 202, whichfirst isolates both signal and processor grounds and then removesfrequencies above the Nyquist limit. In this manner, crosstalk betweendigital and analog signals is reduced. Audio signals must be isolatedfrom digital circuits in order to prevent interaction and crosstalknoises. If not done effectively, these problems propagate throughout theaudio component chain as well as within the encoder electronics. Thesupersonic filter is needed to eliminate high frequency components inthe incoming signal, including radio frequency leakage and other noises,which would otherwise create alias and foldover distortions or beatswhen the signal is sampled. The output of the filter is applied to anoversampling A to D converter. In the embodiment shown in FIG. 15, thesignal is sampled at 4 times the final 44.1 kilohertz frequency. Inanother embodiment, we used a converter running at 8 times 44.1 kHz. Aspart of the transient analysis described below, we are interested infrequencies up to at least 40 kiloHertz, and therefore, the filterresponse begins to role off above this region. In both cases, the cutofffrequency of the filter is well above the normal audio range, so thatthe filter can have a gradual roll off and not introduce audible phasedistortions. The "brick wall" alias filter for the encoded signal isimplemented as a digital filter in the decimation process describedbelow. It is essential, however, that the response of the analog filterbe down below the resolution limits for frequencies which would aliasinto the audible range (i.e. input above 132 kHz for 4-timesoversampling), since these alias products cannot be distinguished fromthe program material or filtered out later.

State of the art filter designs attempt to keep alias and foldovernoises well below the resolution limits of the digital code. The wellknown Compact Disc encoding yields 16 bits of data sampled at 44.1 kHz.Input frequencies above 22.05 kHz exceed the half sampling rate Nyquistlimit and simply will not play back. Instead, one gets lower frequencydifference components which, to be inaudible, should be at least 85 dBdown for a standard CD. However, the invention needs a digital signalwith higher resolution, which means proportionately more stringentfilter characteristics. Since we are dealing with a signal withapproximately 20 bit resolution, as described below, we need to keepinput signals which would cause alias products down by at least 108 dB.Because of their similar non-musical character, crosstalk interferencebetween analog and digital processes must be at least as low.

While analog filtering and isolating operations are functionallyseparate operations, the required circuits are related and often workbest when constructed together from one group of components. Gooddesigns may have fully balanced push-pull signal paths, as well asseparate power, grounding, and shielding.

The output of the supersonic analog filter 202 is applied to the sampleand hold and analog to digital conversion subsystem 203 through asumming junction in which dither is added. The continuous analog signalsare sampled at regular intervals and the sample voltage held unchanginglong enough to be converted to a binary number or word which representsthe amplitude of the sample. As has been discussed previously, fastersampling rates give more points to define the signal waveshape andlonger digital codes or more bits give finer resolution for each sample.Accurate conversion is very difficult and many clever techniques toachieve it are represented in commercial products. We are currentlyusing a commercial hybrid integrated sample and hold and A to Dconverter which can operate at a 176.4 kiloHertz or 4 times oversamplerate and produce 18 bit digital words representing the sampleamplitudes. This unit is at the limits of the current state of the artin commercially available converters. Prior to the availability of theseconverters, we used another commercial converter with 16 bit accuracy atan 8 times oversampling rate.

In order to get resolution in an A to D system which matches thecapability of modern converters, great care must be exercised tominimize noise and analog-digital interaction. One of the techniqueswhich we use is called silent conversion. In order to prevent digitalinterferences to analog signals and conversion timing, the entire logicand conversion system shuts down prior to the critical samplingoperation. Noise from cables, IC's and other parts becomes ten to onehundred times less and a signal sample to accurate to millionth's of avolt occurs. Once the analog signal is sampled and safely held, theconversion process resumes and the digital code is sent to the digitalsignal processors. Other systems do not work like this and are severelyhindered by noise, crosstalk, or glitches.

Another aspect of A to D and D to A conversion which is very important,as discussed earlier, is the minimization of sampling time jitter.Findings recently reported in the audio trade press indicate that ajitter of 100 picoseconds in the sampling time is clearly audible. Inorder to keep this jitter to a minimum, we place the system clock 209 inthe A to D converter module. We use a clock circuit designed to havevery low phase noise, and use a short path to the converter. The clockis also buffered and used to provide the master timing to the rest ofthe system.

Oversampling, in addition to the advantages regarding analog filterdesign previously discussed, allows a given converter to achieve ahigher amplitude resolution, or more bits to represent signal levels,when decimated. Each additional bit doubles the encoded resolution toyield an almost 6 dB greater dynamic range. In 4 times oversampling, forexample, 4 samples are taken for each one present in the final format,and those extra samples contain more information about the originalsignal. Some decimating converters simply discard this additionalinformation, but we convert it into amplitude resolution by usingsubtractive dither. One of the functions of the first DSP subsystem 205is to generate a dither signal, which can take one of several forms,including a sawtooth, a sine wave, and a pseudo-random noise. A processwithin the DSP generates small seemingly random numbers, which arescaled to fractional bit levels. These numbers are applied to a digitalto analog converter 207 whose output is smoothed and scaled orattenuated in 208 to achieve fractional bit levels when added to theincoming analog signal. The voltage is added to the audio signal therebycreating a vernier effect. Within the DSP system 205, the dither numbersare delayed to match the system delay for samples coming from the A to Dconverter 203 and the dither is subtracted out again. When the signal isaveraged by the low pass filter process in 205 as part of decimation,the smallest signal components can be determined to fractions of a leastsignificant bit of the converter. These operations must occur at theincoming sampling frequency and, in the present scheme of 4 timesoversampling, up to an extra two bits of resolution is possible.

The digital output of the analog to digital converter is directed to thesignal analysis subsystem 211 and through delay subsystem 204 to thefirst digital signal processing subsystem 205. The delay provided by 204allows analysis of the signal to be made and a process control decisiontaken before the signal reaches the DSP system. In this way, the DSP isnever "surprised" or caught off guard by changing signal conditions.

In a presently preferred embodiment, the digital signal processingsubsystem 205 is implemented using two commercial DSP processors with 24bit word length and 56 bit accumulators. It performs a variety offunctions, including: generation, delay and subtraction of the dithersignal, described above; low pass filtering the signal using a varietyof filters; decimation of the signal to the industry standard samplingrate; and handling the transitions from one filter to another under thecommand of the process control subsystem 211. First, a delayed copy ofthe dither which was added to the analog signal prior to conversion issubtracted from the incoming digital signal. The signal then undergoesdecimation, which involves low pass filtering followed by repeatedlydiscarding three samples and keeping the fourth. It is this digitallyimplemented low pass filter which performs the anti-alias function forthe signal at its final sampling frequency, and, as discussedpreviously, no single filter implementation can be ideal under allprogram conditions because of the steep transition between the passbandand stopband required. While a symmetrical finite impulse responsedigital filter is free of the variable group delay and phase distortioneffects which plague analog filters, there are still tradeoffs betweenalias rejection, transient response, and passband frequency response.The invention solves this problem by using different filtercharacteristics for different signal conditions, and making a smoothtransition or merge from one filter to another. The implementation ofthe filters is a standard one for FIR filters using multiply andaccumulate functions. The result of decimation is a signal havingapproximately 20 bit resolution at one-times sampling rate. This 20 bitaccuracy necessitates a filter stopband rejection of at least 108 dB tokeep alias products below the resolution of the signal.

The output of DSP subsystem 205 is a digital signal at the industrystandard sampling rate (44.1 kHz for CD's) having 20 bits ofinformation. This signal is passed to the second digital signalprocessing subsystem 210, which packs the 20 bit resolution into 16 bitwords matching the industry standard and adds control information foruse by the reproducer. These operations are carried out under thecommand of the process control subsystem 211. The information packing isaccomplished using a digital implementation of the analog systemdescribed earlier. For level peaks in the program, an instantaneous softlimit transfer function is used. Since it is implemented in an exactmathematical way, the transfer function can be chosen to have minimumaudible effect for undecoded playback and can be exactly reconstructedin the reproduce decoder. It is also possible for the process controlsubsystem 211 to alter the limit parameters, such as changing the limitthreshold in response to the degree of limiting which may already havebeen applied to the signal before it reached the encoder. In doing so,the controller can also send the parameter information to the reproducerusing the control codes hidden in the signal.

For very low level signals, an average gain compression is used toincrease the system gain. This gain increase raises the level of thosesmall signals further into the upper 16 bits of the digital word, andthen the 20 bit word is rounded off to 16 bits, matching the industrystandard format. The gain is controlled by subsystem 211, which looksahead in time from the point of view of the DSP system 210. 211 sees anundelayed signal, while the DSP system gets one delayed by 204 and 205.The control subsystem also inserts control codes which tell thereproducer what it has done with the gain. The second DSP subsystem isalso used to apply "dynamic dither" or noise shaping as discussed abovein the analog description.

The final task of the DSP system 210 is to encrypt and insert thecontrol codes into the least significant bit of the digital words. Thedetails of this process are discussed later. These are the codes whichtell the decoder what has been done to the signal, so that it can carryout complementary processes.

Both DSP subsystems receive commands from the signal analysis andprocess control subsystem 211. This module receives the oversampleddigital signal directly from the A to D converter, conditions it,analyzes it, and based on the analysis, makes process control decisionsand sends commands to the DSP modules. It also generates the controlcodes for the reproducer which are included in the encoder output. Themodule uses digital versions of the analog algorithms discussed earlier:

It uses ratios of high frequency content to total amplitude along withdetected isolated transients to select filter programs for thedecimation filter.

It measures the average signal level of the broad middle frequencyspectrum and uses the results to control the gain of the low levelcompressor. It also generates reproducer control codes to correctlycomplement the encode gain structure.

It measures the average level of low level high frequency signals, andinvokes dynamic dither insertion of extra high frequencies whenappropriate.

It analyzes the distribution of peak amplitudes to determine if theincoming signal has been limited prior to the encoder. If so, it canraise the threshold of the encoder's soft limit function, or turn it offaltogether.

It can compare the decimated signal to the oversampled one delayed tomatch the decimation to look for isolated bursts of high frequencyinformation which represent transients which would not fit within thenormal 22 kilohertz bandwidth. These difference signals can be sent tothe reproducer in the control channel, spread in time, so that thereproducer can correct the transient on playback.

It can also use the transient analysis to control slew rate limiting ofthe main signal as an alternate approach to increasing the apparentbandwidth of the system, as previously discussed.

It controls the insertion of hidden codes in the least significant bitof the encoded signal, putting them in when needed and letting the LSBbe used for the main signal when it is not needed for control.

The process control subsystem is the nerve center of the encoder, makingthe decisions and controlling the functions of the DSP units. It is notnecessary for a given implementation to incorporate all of the featuresabove. For economic reasons, it may be desirable to only include aparticluar subset. Since the encoder uses control codes to tell thereproducer what it is doing, a more capable reproducer will not beconfused, and a less capable one will ignore those functions that itcannot complement.

The digital data output from the second DSP module 210 goes to theformat converter and then to the recorder. Compact Disc, digital audiotape, etc. operate on similar encoding principles. However, thesesystems have different recorded formats and electronic signals for thesame 16 bits of encoded program data. In the industry standard formatconversion subsystem 212, specialized IC chips are configured to addprogram track information and other housekeeping information to the dataand combine the two channels of 16 bit program digital data into asingle data stream which has been configured to industry standardformats. The end result is a combined data and operational code madecompatible with the input of a standard digital recorder. This moduleperforms functions common to all digital recording systems, and usescommercially available special function integrated circuits to performthe format conversions.

As best observed in FIG. 16, a more detailed diagram of the presentlypreferred digital embodiment of the decode system, highly specializedoperations are performed by functional groups of electronic components.In the playback subsystem, the first element of the reproduce chaincould be a video player, CD player, receiver or other equipment. Thesecomponents usually have servos, conversions from specialized standards,buffer memories, and occasionally phase or frequency locked timingsystems to achieve stable continuous playback signals. For example, sucha system could be a CD transport. Each type of digital system requiresits unique unscrambling, patching, and fixing operations to eventuallyextract "error free" program digital data and this is accomplished bystandard circuitry within the player or other device. The output of theplayer is a stream of digital data in one of several industry standardformats, and this stream of data forms the input to our decode system.

Referring to FIG. 16, the data from the reproducer is applied to aformat converter 220, in which one of the industry standard serialdigital data formats is converted into a form suitable for useinternally within the decoder. The data is normally split into right andleft channels at this point for separate processing. This formatconversion is carried out using commercially available integratedcircuits designed for this function. This subsystem also may provideservo feedback control to the transport to control the incoming datarate, and it provides timing information to the decoder system clock.

The data output of the format converter goes to the control decodemodule 221. This subsystem is complementary to the process controlsubsystem 211 in the encoder. Its functions include detecting anddecoding the hidden control codes inserted by the encoder, possible codestripping or removal of the code from the signal, signal analysis of thedata signal, and generation of process control signals to control theDSP modules based on the nature of the signal and the hidden codes.

The data signal then goes to delay module 222, which gives the controldecode module 221 time to figure out what to do with the signal beforeit gets to the first DSP subsystem 223. The first DSP module 223 is thecomplement to module 210 in the encoder. It does a peak expansion whichrestores the peaks limited in 210. It does a low level gain expansion,restoring the low level dynamics compressed in 210. It can complementthe low level forcing of high frequencies in the dynamic ditheroperation, restoring a flat frequency response and lowering quantizationnoise. It performs some housekeeping functions, and its signal outputhas 18 to 20 bits of real information at the one-times sampling rate(44.1 kHz for CD).

This more accurate digital signal at the media sampling rate is routedto the second digital signal processor subsystem 224, which iscomplementary to encode module 205. In this subsystem, the signal isinterpolated to a higher sampling rate using a variety of smoothingfilters which are chosen to complement the decimation filters in 205.

All D to A conversion systems involve a smoothing operation to convertthe discreet sampled signal back to a continuous analog waveform.Digital interpolation is frequently used to increase the sampling rateby calculating a larger number of steps representing the continuouswaveform. A larger number of smaller amplitude steps reduces the burdenplaced on the analog smoothing filter 227 following the conversion backto analog form. Most player circuits employ some version of thistechnique. Again an "oversampling" has occurred. However, in a normalplayer, the information content between input and output frominterpolation has not changed since the filter cannot create newinformation from its curve fitting computations. By contrast, ourinterpolation subsystem has knowledge of the signal resulting from ananalysis made in the encoder prior to the bandwidth limiting decimation.This information has been sent to it through the control channel in theform of filter selection control, and transient correction orenhancement data, and thus this interpolator can restore some of theinformation deleted by the decimator.

The decode system can also provide some improvement to playback ofstandard unencoded signals by analyzing the incoming signalcharacteristics in module 221 and using the results to pick a smoothingfilter which is probably best. This single ended operation results in animprovement over a conventional player, but it cannot achieve theperformance of the full system.

The smoothing or interpolation filters in DSP subsystem 224 are standardfinite impulse response or FIR types, which are made symmetrical toavoid phase distortions. The subsystem must implement smooth transitionsor merge operations from one interpolation filter to another in the samemanner as the decimator does. It may also include transient synthesisand slew rate modification, similar to the analog implementationdiscussed earlier.

In summary, the first decoder DSP module 223 restores amplituderesolution, and the second DSP module 224 restores frequency ortransient resolution. Both of these operations are the complements ofoperations in the encoder.

The high resolution oversampled signal goes to the digital to analogconverter subsystem 225. As in the encoder, we use a commerciallyavailable D to A converter module which represents the current state ofthe art. The current embodiment uses 20 bit converters operating at 4times oversampling. We have also used 18 bit converters at 8 timesoversampling. As in the encoder, great care must be taken to isolate theanalog signal from the digital noise, and sample timing jitter isminimized by using a low noise master clock tightly coupled to theconverter module. The analog output goes to the supersonic filter.

In the analog smoothing filter subsystem 227 and output buffer amplifiersubsystem 228, final rounding and removal of supersonic signals occurswith an analog low pass filter along with amplification to standard linelevels and output impedance. A sophisticated design such as ours treatsthe analog and digital filtering as a whole system to achieve thebenefits both methods offer. As in the encoder, the isolation of digitaland analog processes is achieved through fully balanced digital andanalog systems, floating power supplies, and isolated grounding schemeswhich prevent interaction with cables and other external components. Theresult is a line level analog output signal. This completes thedescription of the signal path from the analog input of the encoder tothe analog output of the decoder.

The following is a description of the control channel in the presentlypreferred embodiment which allows us to send control commands andauxiliary signal information from the encoder to the decoder in the samesignal as the main program.

The command codes and other auxiliary data are encrypted with apseudo-random noise and inserted into the least significant bit of themain signal digital words in a serial fashion, one bit per word. The LSBof the audio is replaced by a "random" noise for the duration of thecontrol insertion. (Of course, more than one bit could be "borrowed" forthis purpose, but more of the main program would be lost.) The system isset up so that when the control channel is not needed, the LSB carriesthe normal audio signal. Since the digital to analog converters in mostof the current generation of digital audio products are not accurate to16 bits; the loss of the 16th bit will not be audible during undecodedplayback, as long as the information inserted there has noise-likeproperties. Even in high quality systems which do resolve all 16 bits,the insertion is not normally audible because the LSB of most programsalready has very noise-like properties. The low level gain compressionand dynamic dither described previously raise the level of the programduring very quiet periods and help hide the code insertions during thoseprogram conditions under which they might be noticeable. In typicalclassical music programming, the control signal would be inserted forintervals of about a millisecond each occurring several times per secondat most. The loss of full program resolution for these brief intervalsis not noticeable.

Circuits to create random noise, modulate a control signal, insert it inthe LSB of the data stream and then retrieve and decode it have beenassembled and made to initiate filter selection from a hidden controlsignal. These circuits are included in Appendix B.

The process control signal is hidden in the least significant bit of thedigital audio channel by modulating it with a noise signal. Our circuitconsists of a pseudo-random noise generator based on a shift registerwith feedback which implements a maximal length sequence. This type ofgenerator produces a deterministic sequence of bits which sounds veryrandom, and yet is a reproducible sequence. The output of the noisegenerator is added to the control signal modulo-two (exclusive-or'ed),modulating the signal with noise, or scrambling it. The result is theninserted into the least significant bit of the record serial datastream. On the play side, the least significant bit is extracted fromthe serial digital stream and the output of a matching shift register issubtracted from it, modulo-two (exclusive-or again). The result is theprocess control signal, unscrambled again.

There are two basic variations of this scheme. The first version usestwo noise generators, one on the record side and one on the play side.The record generator noise is added to the signal, and the playgenerator noise is subtracted. If the two generators produce the samebit sequence, the original signal is recovered. The problem is that theplay generator must be synchronized with the noise sequence added duringrecord. While there are many well-known approaches to solving thisproblem which are covered in the literature on spread spectrumcommunications, it is still a non-trivial problem. Although thisapproach is feasible, a presently preferred embodiment of the systememploys the following technology.

In the preferred embodiment, the sum of the process control signal andthe generator output is fed back to the generator input. Thiseffectively "folds" the signal into the generator sequence so that thescrambled signal depends only on the recent history of the bits, and theplay side contains a matching shift register with no feedback. Becausethe play side only uses "feed-forward" addition of the bits in the shiftregister, it is guaranteed to become synchronized as soon as N+1 bitshave arrived, where N is the length of the shift register. Thedisadvantage of this approach is that it is possible for the noisegenerator to become stuck temporarily, depending on the characteristicsof the process control signal. The probability of this happening can bemade arbitrarily small by increasing the length of the shift registers.In the implementation shown in Appendix B, which uses a 17 bit shiftregister, the probability is on the order of 1 in 100,000 that a bitsequence might occur which could cause the generator to stick. By goingto a 31 bit shift register, the probability drops to about 1 in 2billion, which corresponds to once every 12.6 hours for a CD. If theprocess control signal is changing rapidly, the artifact of a "stuck"noise generator will be of short enough duration to be noise-like andinaudible. The problem of a stuck noise generator is only relevant ifthe control sequence is inserted continuously. In the preferredembodiment, in which the control is only inserted for brief intervals,it is not a problem for two reasons. First, since the LSB is returned tothe program signal most of the time, the stuck generator output is notinserted into the signal. Secondly, dynamic insertion requires the useof a synchronizing sequence, as described below, which can be designedto guarantee that the generator does not get stuck.

Dynamic insertion of the control signal into the LSB or sharing the LSBwith the main program data means that the reproducer has to be able toidentify the commands embedded within the stream of arbitrary mainprogram data. This is accomplished by preceding a command code with asynchronizing sequence of bits which the decoder looks for in the datastream. The sequence can be made long enough that the probability of itsoccurrence in the program data is extremely small, assuming that theprogram data is essentially random. Of course, one must avoid patternsthat might appear with more than random frequency, such as long stringsof ones or zeros which could occur during silent periods in the program.False triggering of the reproducer on program data can be completelyeliminated on recordings incorporating the invention by having theencoder monitor the program data stream during recording and alter theleast significant bit in one word if the synchronizing sequence is aboutto occur, thereby preventing it. This results in a bit error probabilityon the same order as the prevented false trigger probability, which canbe made much less disruptive than the insertion of control codes, and soinconsequential.

It should be noted, that the aforedescribed technique can be used tohide arbitrary digital data in a digital audio signal or other digitalsignal representing analog data in which accuracy to the leastsignificant bit is not continuously necessary. Such data inserted inplace of or in addition to our process control signal could be used tocontrol a multi-media presentation or for some other completelyunrelated purpose.

It will be apparent from the foregoing description that those ofordinary skill in the audio, digital and data processing arts should beable to utilize a wide variety of computer and other electronicimplementations in both hardware and software to practice many of theanalysis, evaluation, encoding, decoding and compensation techniquesembodied within the methods and apparatus of the present invention.

The aforedescribed systems of the present invention satisfy a longexisting need in the art by providing new and improved digitalencoding/decoding methods and apparatus for ultra low distortionreproduction of analog signals and which are also compatible withindustry standardized signal playback apparatus not incorporating thedecoding features of the present invention. In addition, signals lackingthe encoding process features of the invention are likewise compatiblewith playback decoders which do embody the invention and are providedsome overall enhancement.

The present invention provides an improved encode/decode system enablinga predetermined balance or interplay of gain, slew and wave synthesisoperations to reduce signal distortions and improve apparent resolution.Analysis is made of waveform characteristics during the encodingprocess, and the results of this analysis are subsequently utilized inthe decoding process to more accurately reconstruct the originalwaveform, while minimizing the deleterious effects normally encounteredin sampling and converting analog signals to digital signals andsubsequently reconverting the digital signals back to an accuratesimulation of the original analog waveform.

In accordance with the invention, control information developed duringthe aforedescribed waveform analysis is concealed within a standarddigital code and this information is subsequently used to dynamicallychange and control the reproduction process for best performance. Theseconcealed control codes trigger appropriate decoding signalreconstruction compensation complementing the encoding process resultingfrom the signal analysis. Since the control code is silent and theoverall digital information rate is normally fixed, the process canoperate compatibly with existing equipment and to manufacturer'sspecifications and standards. In addition, and as previously indicated,signals lacking the encoding process features of the invention arelikewise compatible with playback decoders which do embody the inventionand are afforded some beneficial enhancement.

It will be apparent from the foregoing that, while particular forms andseveral aspects of the invention have been illustrated and described,various modifications can be made without departing from the spirit andscope of the invention. Accordingly, it is not intended that theinvention be limited, except as by the appended claims. ##SPC1##

We claim:
 1. In a data processing system, a method for converting andencoding an analog waveform to a digital format, comprising the stepsof:interrupting all system operations to create a period of electricalsilence; performing digital sampling of said analog waveform during saidperiod of electrical silence; and resuming system operations after saiddigital sampling has been completed.
 2. In a data processing system, asilent system, comprising:data logic and conversion means; data samplingmeans; and means for shutting down operation of said data logic andconversion means during operation of said data sampling means.